Ceramic antenna module and methods of manufacture thereof

ABSTRACT

Antenna modules that contain composite meta-material dielectric bodies that have high effective values of real permittivity and reductions in physical lengths of electrically conducting elements.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of the priority of U.S. patentapplication Ser. No. 14/089,465, filed Nov. 25, 2013, which in turnclaims priority of U.S. patent application Ser. No. 13/471,012, filedMay 14, 2012, which in turn claims priority of U.S. patent applicationSer. No. 12/177,002, filed Jul. 21, 2008, which in turn claims priorityof U.S. patent application Ser. No. 11/243,422, filed Oct. 3, 2005,which in turn claims the priority of U.S. Provisional Patent ApplicationNos. 60/615,174, filed Oct. 1, 2004, and 60/716,306 filed Sep. 12, 2005,and the contents of all of which applications are incorporated herein byreference as if set forth in their entirety.

FIELD OF THE INVENTION

The present invention relates generally to the construction of ceramicdielectric meta-material components that have high effective values ofreal permittivity but minimal body losses, and, more particularly, tosmall form factor wireless circuit modules operating in the radiofrequency (RF) spectrum that contain at least one antenna element or atleast one ultra-low loss transmission line.

BACKGROUND OF THE INVENTION

As used in the descriptions that follow, the term “meta-materials”refers to materials that possess unique macroscopic properties due tofiner scale repetition or alterations of one or more secondary materialswithin the host material to alter the bulk body's dielectric orconductive properties.

“Electromagnetic Band-Gap” (EBG) materials, also known to those skilledin the art as “Photonic Band-Gap” materials, are meta-materials thatcontain one or more secondary phase dielectric inclusions that areorganized in periodic array(s) with periodic spacing(s) havingdimensions that are an appreciable amount of a center frequency'swavelength so as to cause constructive and destructive interference overa particular range of electromagnetic frequencies. An EBG materialeffectively attenuates frequencies that fall within in its bandgap orstopband as the periodic dielectric inclusions inhibit propagation dueto the destructive interference of reflections off the periodic array.

The term “Perfect Electrical Conductor” (PEC) refers to an infiniteconductive surface that causes electric field components of anelectromagnetic wave incident upon the PEC to be totally reflected 180°out of phase with the incident wave.

The term “Perfect Magnetic Conductor” (PMC) refers to an imaginarysurface generated by a periodic array of coupled inductor and capacitorelements that causes the electric field components of an electromagneticwave incident upon the PMC to be totally reflected completely in phasewith the incident wave. A finite dimension imperfect PMC can bepractically constructed using meta-material construction techniquesusing dielectric material inclusions into a dielectric host that hassuitable permittivity and permeability values and periodicity tosimulate the coupled inductor and capacitor elements of the imaginaryPMC. A PMC may alternatively be known as an artificial magneticconductor (AMC).

The term “EGB Defect Resonator” refers to a resonant structure formed byselectively removing or eliminating one or more dielectric inclusionelement(s) from the periodic array that forms the EGB meta-material in amanner that permits waves within a narrow frequency band of the EGB'sbandgap or stopband to propagate freely through the medium and/or to belocalized within a specific region of the meta-material dielectricdefined by the defect.

RF electronic modules typically comprise one (or more) semiconductorchip(s) that is (are) connected to passive circuit elements (resistors,inductors, and capacitors) and/or other discrete circuit elements suchas diodes, transistor switches, SAW filters, baluns, or impedancematching networks, among others, through a passive interconnectstructure. The passive interconnect structure is formed by routingelectrical signals over conductor leads that are attached to the surfaceof an organic or ceramic dielectric layer or are embedded within saidorganic or ceramic dielectric layer. RF electronic modules represent thenext generation of microelectronic integration in that they integratethe semiconductor die with additional components that cannot bemanufactured integrally to the semiconductor IC as a single part.Modules are gaining popularity in large market applications because theyreduce part count and conversion costs to the OEM. In cost sensitiveproduct applications, modules having an interconnect structure that isformed with organic dielectric are preferred; however, in applicationsthat utilize signals operating at high frequency (e.g., f≧1 GHz) or thatare subject to high thermal loads, more expensive ceramic dielectricsmay be used to reduce absorptive dielectric loss and boost signalintegrity. Therefore, a module containing a passive ceramic interconnectstructure that offers low-cost and improved signal integrity has greatvalue in wireless circuit applications.

Wireless circuits are used to form a network connection between a mobileplatform (such as a cell phone or laptop computer) and are findingincreasing application in fixed local area networks as well as radarsystems, due to their low cost and easy installation. Radiocommunications are managed through the device's RF front-end, which willtypically operate near to or at GHz signaling frequencies where theinsertion loss of passive circuit components is widely known toincrease.

The power budget is often a prime concern in mobile platform design, somethods that minimize insertion loss among components used to assemblethe front-end have great value. For instance, the front-end of a CDMAcell phone may typically contain a power amplifier (PA) die, a duplexerswitch that alternately modulates transmit and receive modes to theantenna, an isolator circuit, and a switch/diplexer component thatdiscriminates individual frequencies of interest. This front-endcircuitry will typically impose roughly 4 dB of signal loss, so mosthigh power PA die are designed to accommodate 4 dB of loss between PA'soutput port and the antenna's feed point. Therefore, methods that canreduce this loss would extend battery life by allowing the power budgetdedicated to radio broadcast to operate significantly longer than aconventional front-end circuit.

Signal loss is minimized in the module by limiting the overall lengththe signal must travel between the PA and the antenna, decreasing thesignal loss per unit length along the transmission line(s) used todirect the signal from the PA to the antenna, minimizing the number ofloss generating components that are needed to process the signal(s)along that path, and by ensuring excellent impedance match along theentire path between the signal source (the PA) and the antenna so as tominimize reflected energy along that path.

Physical size is another critical issue in mobile platform design.Today's state of the art has reduced the size of a cell phone's frontend to an area that is roughly 1.5 inch². Therefore, an invention thatwould allow the entire function of an RF front end to be reduced in sizeto an area that is roughly the size of a PA die (4 mm×4 mm or 0.02 inch)has great value.

Physical size requirements are greater in interconnect structures thatpermit semiconductor dies to be flip-chip mounted on the interconnectstructure. Input/output (I/O) contact pads 101 such as shown in FIG. 1are mounted on the surface of the dielectric 111 in which conventionalinterconnect structures 103 are embedded, causing wells 105 to be formedbetween the pads 101 protruding from the dielectric's surface. Soldermasks 107 are used to prevent solder balls 109 that connect theinterconnect structure 103 to the semiconductor die 113 from wickinginto the wells 105 during reflow processing and forming short circuitsbetween adjacent pads. Recent increases in semiconductor integrationhave produced I/O pad pitches that are beyond solder mask manufacturingtolerances. This forces many module manufacturers to backside mount thesemiconductor die and connect it to the interconnect structure throughwire bonds located on the die's periphery, which then occupies a largerfootprint. Therefore, modules that have I/O pads embedded within thedielectric of the interconnect structure, thereby allowing high-densitysemiconductor die to be flip-chip mounted within a wireless circuit aresignificant and have great value.

U.S. Pat. No. 6,027,826 to de Rochemont, et al., disclose articles andmethods to form oxide ceramic on metal substrates to form laminate,filament and wire metal-ceramic composite structures using liquidaerosol spray techniques. U.S. Pat. Nos. 6,323,549 and 6,742,249 to deRochemont, et al., disclose articles that comprise, and methods toconstruct, an interconnect structure that electrically contacts asemiconductor chip to a larger system using at least on discrete wirethat is embedded in silica ceramic, as well as methods to embed passivecomponents within said interconnect structure. U.S. Pat. Nos. 5,707,715and 6,143,432 to de Rochemont, et al., disclose articles and methods torelieve thermally-induced mechanical stress in metal-ceramic circuitboards and metal-ceramic and ceramic-ceramic composite structures. Thecontents of each of these references are incorporated herein byreference as if laid out in their entirety.

SUMMARY OF THE INVENTION

The present invention provides circuit modules that contain at least oneantenna element or at least one ultra-low loss transmission line in aninterconnect structure supplying RF electromagnetic energy to an antennaelement formed by embedding discrete wire or other type conductorswithin a pure amorphous silica, alumina, titania, or other suitableceramic dielectric. The present invention also provides a method ofconstructing a ceramic dielectric meta-material that has a higheffective value of real permittivity but which minimizes reflectivelosses, through the use of a host dielectric (organic or ceramic) havinga relative permittivity substantially less than the embedded ceramicdielectric inclusions used to form the dielectric meta-material, andfurther allows the physical lengths of the antenna element(s), formedeither with discrete wire conductors or patterned film, to be minimizedwithout adversely impacting radiation efficiency. The meta-materialstructure may additionally provide frequency band filtering functionsthat would normally be provided by other components typically found inan RF front-end, and methods to construct the same. In a preferredembodiment of the invention the host dielectric is selected to have aminimal loss tangent to maximize the efficiency of the circuit module.

The needs for the invention set forth above as well as further and otherneeds and advantages of the present invention are achieved by theembodiments of the invention described herein below.

In one aspect, the present invention provides a method of constructing asemiconductor module such as, for example, an antenna module, thatcomprises forming a meta-material dielectric region in which highpermittivity (high-κ) dielectric ceramic is incorporated within a hostdielectric medium, preferably an ultra-low loss amorphous silica,alumina, titania or other suitable ceramic dielectric host to enhancethe real permittivity while minimizing the dielectric loss of themeta-material dielectric. The embedded high-κ region is dimensioned,sized and/or otherwise adapted for connection to electrical componentsof the semiconductor module. In certain embodiments includes a composite(meta-material) dielectric body formed of one or more ceramic dielectricinclusions composed of materials having a relative dielectricpermittivity ≧10 and/or a relative dielectric permeability ≠1 embeddedin a dielectric host material comprised of amorphous silica, titania,tantalates, pure alumina, and admixtures thereof, and/or of an organicmedia. The combination of host material and inclusions impart upon thecomposite dielectric body an effective permittivity ≧4. The antennamodule also includes a ground plane and at least one contact pad in ametallization layer adjacent the composite dielectric body, and one ormore electrically conductive elements, such as antenna elements that,due to the presence of the composite dielectric body is/are resonantover a band of frequencies and that have a length (or lengths)≦50% ofthe length that would be required to maintain the same resonance withoutthe composite dielectric body. Controlling the grain size of theinclusions, as described below, adds stability to the antenna module.

In another aspect, the present invention provides an antenna moduleconnectable to at least one semiconductor die and methods forconstructing the module that includes an AMC ground plane. The AMCground plane structure includes the composite (meta-material) dielectricbody, an electrical ground plane and contact pads (through whichelectrical connection to one or more antenna elements can be made to oneor more corresponding semiconductor die), and a periodic array ofcapacitor pads disposed at a surface of the composite dielectric bodythat are electrically shorted through the composite dielectric body. Thearray has substantially uniform periodicity and spacing. A reflected Efield component of an electromagnetic wave incident upon the AMC groundplane structure that has a frequency within a bandgap range offrequencies will experience an induced phase shift that produces acondition of constructive interference between the reflected E fieldcomponent and E-field components emanating from the at least oneelectrically conductive element that substantially share the samedirectional propagation vectors as the reflected E field component.

In another aspect, the present invention provides novel low-lossinterconnect structures useful in any type of circuit module thatrequires a connection between a semiconductor die and a microelectroniccomponent. The interconnect structure includes a dielectric body inwhich one or more transmission lines and electrical connectors areembedded (entirely within or at the surface of the dielectric body.) Thedielectric body is comprised of multiple layers of a host dielectricmaterial having a relative dielectric permittivity ≦4.5 or,alternatively, having a loss tangent ≦3×10⁻³. In certain embodiments,impedance matching among the various electronic components (includingcomponents formed of discrete wire) is achieved through use of a layerof a high-κ ceramic material having a relative permittivity ∈_(R)≧25 inthe module and dimensioning the electrically conducting elements to havesubstantially similar cross-sectional areas and forms.

In another aspect, the present invention provides a meta-materialdielectric body, useful in a variety of circuit modules, and a method ofconstructing same. The meta-material dielectric body may be formed byembedding one or more dielectric inclusions in a host material bodyhaving a loss tangent that is ≦3× 10⁻³, resulting in a composite(meta-material) dielectric body having an effective relativepermittivity ≧4. As used herein, the term “embedding” does not connoteany particular order of operation, as will be clarified below. Embeddinga high-μ material within an ultra-low loss amorphous silica or alow-loss titania host enhances the real permeability while minimizingthe magnetic losses of the meta-material dielectric. In preferredembodiments, the maximum nominal grain size is controlled to be ≦50nanometers, or preferably ≦35 nanometers, which impart upon themeta-material dielectric body an effective permittivity that changes asa function of temperature ≦5×10⁻²° C.⁻¹ (for the be ≦50 nanometersembodiment) and ≦8×10⁻³° C.⁻¹ for the latter embodiment. The hostmaterial may be composed of amorphous silica, titania, an organicmaterial such as Roger Duroid, GTEK and Teflon (PFTE) organic polymers,admixtures thereof, and admixtures of amorphous silica (α-SiO₂) andeither titania (TiO₂) or alumina (Al₂O₃) having a corresponding chemicalcomposition Si_(1-x)Y_(x)O₂, where 0≦x≦1 and Y is Ti or Al. Othercompositions are described below.

In another aspect, the present invention provides circuit modulesincluding one or more electrically conductive elements (e.g., antennaelements or transmission lines) disposed parallel to a metallizationlayer including a ground plane and at least one contact pad. The circuitmodules include a composite dielectric body adjacent the metallizationlayer on the same side of the ground plane on which the conductiveelements are located. The composite dielectric body includes at leastone ceramic dielectric inclusion embedded in a host material having arelative dielectric permittivity ≦4.5. In alternative embodiments, thehost material has a loss tangent that is ≦3×10⁻³, and the compositedielectric body has an effective relative permittivity ≧4.

In a preferred embodiment, the present invention provides afrequency-selective antenna (FSA) element and method of manufacturethereof by embedding at least one narrow diameter discrete wireconductor as the electrically conducting elemental within an ultra-lowloss meta-material dielectric.

In another embodiment, the present invention provides terminatingcapacitor pads to further reduce the length and simultaneously managethe manufacturing tolerances on wire bond length, without adverselyimpacting dipole arm length of a miniaturized, frequency-selectiveantenna element embedded in a meta-material dielectric.

In another embodiment, the present invention provides a miniaturizedantenna element and method of manufacture thereof using an aerosol sprayto dispose at least one electrical conducting element relative to (e.g.,embedded within or at the surface of) a composite (meta-material)dielectric body comprised of high-κ ceramic microscopic regions embeddedwithin an amorphous silica, alumina, titania, or other suitabledielectric host, including an organic dielectric.

At least one frequency-selective antenna (FSA) or miniaturized antennaelement(s) may be configured over a conducting ground plane that shieldsat least one power amplifier (PA) die situated on the opposite side ofsaid conducting ground plane from electromagnetic emanations of theradiating antenna element(s).

In a further embodiment, the present invention provides a semiconductormodule having at least one frequency-selective or miniaturized antennaelement disposed on one side of an artificial magnetic conductor (AMC)meta-material structure that shields at least one PA die situated on theopposite side of the AMC from electromagnetic emanations of theradiating frequency-selective or miniaturized antenna element, thatprovides positive gain to the antenna and which suppresses surface-wavemodes that would couple energy radiated by the antenna element(s) toadditional components situated nearby and vice-versa.

The stop-band gap of the AMC may be maximized by incorporating lowpermittivity amorphous silica and/or ferrite materials derived from anaerosol spray into the AMC meta-material structure.

In still another embodiment, the present invention provides asemiconductor module having high permittivity (high-κ) or highpermeability (high-μ) dielectric regions disposed within a hostdielectric material at a periodic spacing and with physical dimensionsthat combine to create an EGB defect resonator having a pass band thatis tuned to the radiation bandwidth of the antenna element(s)incorporated into said host dielectric material that shields theantenna(s) from spurious signal interference.

In yet another aspect, the present invention provides a means to reduceloss by tuning the input impedance at least one antenna element embeddedwithin an amorphous silica host dielectric so that it is matched to the2 Ω, 5Ω or 10Ω output impedance of at least one PA die that iselectrically connected to the antenna element.

In another aspect, the present invention provides a low insertion lossfeed network (interconnect structure) and manufacturing method thereofthat minimizes electrical reflections and signal attenuation within theportion of a semiconductor module that electrically connects at leastone PA die to at least one antenna element through discrete wiretransmission lines embedded in an ultra-low loss amorphous silicadielectric material. Amorphous silica or admixtures thereof maintainsufficient optical transparency to apply automated optical inspectionand indexing that permits formation of vias and contact pad structuresthat are nominally the same physical size as the diameter of conductingelements used in the discrete wire transmission lines.

Contact pads may be embedded within the surface of the ceramicdielectric to allow the PA (or other semiconductor) die to be flip-chipmounted to the low insertion loss feed network without the need for asolder mask.

The PA (or other semiconductor) die may be flip chip mounted to themodule's low insertion loss feed network to permit thermal heat sinks tobe attached to the back side of the PA (or other semiconductor) die.

The present invention also provides a ceramic module and a method forconstructing a ceramic module comprising at least one PA die and havingsingle or multiple inputs and outputs and that is electrically connectedto at least one narrow band (high-Q) frequency selective antennaelement, the narrow band defined as have a bandwidth of less than 3%,preferably less than 0.5%. The frequency selective antenna element isspecifically tuned to match the specific desired frequency band of theinput(s) and output(s), such that the module performs the same functionof an RF front end by using the individual antenna element(s) to isolatea specific frequency and direct its signal(s) to the appropriateinput/output, effectively replacing the collective function of multiplecomponents (SAW filters, switches, diplexers, isolators, pre-amplifiers,etc.) used in the construction of a conventional RF front end.

In a particular embodiment the ceramic module includes at least one pairof high-Q frequency selective antenna elements, with one of the antennaelements having a center frequency tuned to the receive channel of aparticular communications band and the other antenna element having acenter frequency tuned to the transmit channel of the particularcommunications band, and the respective bandwidths of the antennaelements tuned to provide sufficient signal isolation between them, suchthat signals transmitted do not interfere with the signals received bythe other.

In yet another aspect, the present invention provides RF front-endcircuits for a wireless communication devices, such as cell phones, orwireless network access points, including the novel antenna and circuitmodules described above, plus additional electronics as appropriate andas described below.

An important aspect of this invention is the ability to construct high-Qtransmission lines used to form antenna or interconnection networks thatmaintain frequency pass band stability over a wide range of operatingtemperatures.

The invention also advantageously improves radiation efficiencies byreducing internal reflections within the semiconductor module structure.

BRIEF DESCRIPTION OF THE FIGURES

For a better understanding of the present invention, together with otherand further aspects thereof, reference is made to the followingdescription taken in conjunction with the accompanying figures of thedrawing, wherein:

FIG. 1 is an illustration of represents prior art modules wherein thecontact pads used to connect flip-chip mounted semiconductor die to theinterconnect structure that protrude from the feed network's dielectricsurface;

FIGS. 2A-2E are top and cross-sectional views of a fully assembledmodule containing a power amplifier die that is electrically connectedthrough an ultra-low loss feed network (detailed in FIGS. 2C-2E) to twofolded dipole antennae on one side of the die and a thermal heat sink onthe other side of the die;

FIG. 3 represents termination pad geometries to manage manufacturingtolerances in wire bond length without significantly altering capacitiveloads;

FIG. 4 is a meta-material dielectric body comprising secondary phasematerial regions embedded within an amorphous silica host;

FIGS. 5A-5J are illustrations of sequential steps used to create adiscrete wire antenna embedded with a dielectric or meta-materialdielectric body and methods that can be used to match the antennaelements impedance to a semiconductor die;

FIG. 6A-C are cross-sectional illustrations of various antenna modulesthat comprise antenna elements formed with conductive metallizationother than discrete wire;

FIGS. 7A-7B are illustrations showing the use of anti-reflective caplayers used to improve EM transmission between the antenna module and anambient dielectric medium;

FIGS. 8A-8D represent various non-limiting antenna elements other thandiscrete wire that can be incorporated into the antenna module toachieve specific bandwidth design criteria;

FIGS. 9A-9B represent artificial magnetic conductor (AMC) structurescomprising amorphous silica or titania dielectric host materials thatare used to boost gain in low-impedance antenna modules;

FIG. 10 is a graph illustrating the representative phase shift versusfrequency that is characteristic of an electromagnetic wave reflectedfrom the surface of an artificial magnetic conductor;

FIG. 11 is a graph illustrating the characteristic band gap as afunction of frequency that suppresses electromagnetic surface modes frompropagating along the surface of an artificial magnetic conductor;

FIGS. 12A-12E are illustrations showing method steps involved inconstructing an antenna element comprising an artificial magneticconductor comprising ultra-low loss amorphous silica or titaniadielectric host material as a ground plane to an impedance-matchedantenna module;

FIG. 13A-B is an illustration of a meta-material body comprisingultra-low loss silica or titania as a dielectric host constructed as anelectromagnetic band gap (EBG) material that satisfies conditions forBragg scattering;

FIG. 14 is a graph illustrating the electromagnetic band gapcharacteristics of a meta-material body structured to satisfy conditionsof Bragg scattering and the optimal filtering characteristics of a EBGmaterial created from such a body;

FIG. 15 is an illustration showing the use of electromagnetic band gapmeta-materials to isolate the antenna element from noise sources bysuppressing spurious signals and ground loops;

FIG. 16 is an illustration showing the use of electromagnetic band gapmaterials in conjunction with a dielectric body of an antenna module ina manner that is used to isolate or filter electromagnetic signalfrequencies of antenna elements positioned on the surface of or withinsaid EBG dielectric body;

FIG. 17 is a top view of EM field localization around an antenna elementthat is electrically isolated from spurious signals at frequencies notof interest to transmission or reception in the antenna environment bylocated said antenna element at the center of a defect resonator;

FIG. 18 is an illustration of characteristic contours in the attenuationpattern of electric field intensity around an inclusion-free zone of adefect resonator;

FIG. 19 is an illustration a dipole antenna located within theinclusion-free zone of a defect resonator;

FIG. 20 is an illustration of a folded dipole antenna located within theinclusion-free zone of a defect resonator;

FIG. 21 is an illustration of a spiral dipole antenna located within theinclusion-free zone of a defect resonator;

FIG. 22 is an illustration of an antenna module comprising a poweramplifier (or other semiconductor) die attached to an antenna structure;

FIG. 23A-E are illustrations of an antenna module, and methods to makesame, comprising a power amplifier (or other semiconductor) die attachedto an antenna structure through an interconnecting feed network;

FIG. 24 is an illustration of an interconnect structure incorporating anAMC ground plane;

FIGS. 25A-25D are illustrations showing the construction of a low-lossinterconnecting feed network as an independent body separate from thepower amplifier (or other semiconductor) die and the antenna structureor other electrical component that is used to electrically connect andmechanically decouple said power amplifier (or other semiconductor) dieto said antenna structure or other electrical component;

FIG. 26 is an illustration showing the use of an interconnecting feednetwork integrated within an antenna module that electrically connectsto a power amplifier (or other semiconductor) die or other electricalcomponent;

FIGS. 27A,B are illustrations of an interconnecting feed network andantenna structure that is assembled directly on the surface of the poweramplifier (or other semiconductor) die;

FIG. 28 is an illustration of a multiple frequency RF Front-Endarchitecture representative of the prior art that incorporates a networkof filters and switches to isolate and manage signal communications atspecific frequencies of interest;

FIGS. 29A-C are illustrations of the use frequency-selective antennamodules used to eliminate the need for diplexers from a multiplefrequency RF Front-End architecture;

FIG. 30 is an illustration of a multiple-frequency RF Front-Endarchitecture that uses frequency-selective antennas to reduce thereceive-side and the transmit-side each to a single amplifier connectedto an array of frequency-selective antennas each having its bandwidthtuned to a specific frequency of interest;

FIG. 31 is an illustration of a multiple-frequency RF Front-Endarchitecture that uses frequency-selective antennas to reduce thereceive-side and the transmit-side each to a single amplifier connectedto an array of frequency-selective antennas each having its bandwidthtuned to a specific frequency of interest;

FIG. 32 is a plot depicting undesirable shifts in the center frequency,f_(o), of a high-Q antenna that may result from changes in the ambientoperating temperature;

FIG. 33 is a chart showing dielectric response versus ambienttemperature in BST ceramic as a function of the size of the grainscomprising the BST ceramic;

FIG. 34 is an illustration depicting antenna structures used in a smallform-factor CPE radio used to rebroadcast an operator's transmissionwithin the customer's premesis.

FIG. 35A-B are illustrations depicting the use of antenna modules tosuppress wireless network security intrusions from threats beyond asecurity perimeter;

FIG. 36A-B are illustrations depicting the use of antenna modules withina laptop computer;

FIG. 37A-B are illustrations depicting the use of antenna modules withina mobile handset;

FIG. 38 is an illustration depicting the use of antenna modules within awireless device; and

FIG. 39 is an illustration depicting a wireless appliance having antennamodules in accordance with the present invention attached to itssurfaces.

DETAILED DESCRIPTION OF THE INVENTION

Reference is now made to FIGS. 2A thru 4 and FIGS. 6 thru 11, whichillustrate preferred embodiments of various aspects of a wirelesscircuit module 115 in accordance with the present invention, and FIGS.5A-5J and 12A,B, which illustrate the processing steps used to createthe circuit module. Circuit module 115 includes, in its most completeembodiment, an antenna module 114, which is in turn comprised of acomposite ceramic dielectric body, at least one electrically conductiveelement, a low attenuation interconnect structure, and a metallizationlayer adjacent the composite dielectric body and including a groundplane structure and at least one contact pad. References to “discretewires” are intended to denote pre-fabricated, conducting wire bodiesthat are geometrically uniform along their lengths and, when comprisedof metal, which contain significantly higher metallurgical purity andgeometric uniformity than is possible for traces made from conductivepastes or patterned thin films. “Round wire” is understood to definewire structures that have substantially circular cross-sectional area.In contrast, “ribbon” wire is intended to refer to wires having aroughly square or rectangular cross-sectional form. In principal,however, discrete wire can assume any cross-sectional geometry(cross-shaped, hexagonal, star-shaped), without limitation, thatimproves the efficiency of the design. Round geometries will possesshigher self-inductance and, therefore, are preferred in transmissionline structures and antenna element configurations. Ribbon wires havelower self-inductance and are more useful in low-loss (low-impedance)transmission lines. FIG. 2A shows two arms 117A, 117B of a discrete wiredipole antenna (with arms folded in this instance) that are electricallyconnected at feed points 119A, 119B to an amplifier (power amplifier(PA) or low noise amplifier (LNA)) or other semiconductor die 135located on the opposite side of the antenna module 114 and whichterminate on respective capacitive pads 121A, 121B. A second dipoleantenna element 123A,B having its own feed points 125A, 125B andterminating capacitors 127A, 127B is also shown. Antenna elements 117and 123 are constructed over a ground plane 133 that is perforated atthe feed points, through which electrical contact is made to at leastone mounted power amplifier die 135 through an electrical interconnectstructure 137. Although the semiconductor die 135 may be mounted in avariety of ways, flip-chip mounting is preferred in order to facilitatethermal management using a thermal reservoir device 139 providingtemperature stability through the backside 136 of PA die 135. In aflip-chip configuration, the semiconductor die 135 would be mountedopposite the electrical feed network 137 that establishes electricallyconnection at the feed points 119,125 of the antenna elements andcircuits board structures 140 that electrically connect the wirelesscircuit module 115 to a larger system. Conventional interconnectstructures include impedance matching networks to compensate for themismatch between the source impedance of the semiconductor die,typically 2-10Ω, and the load impedance of the antenna element,typically 50Ω. The antenna module's efficiency may be improved byimpedance matching the antenna load to the 2-10Ω source impedance of thesemiconductor die. This impedance match should be maintained throughoutinterconnect structure 137 and antenna feed points. To achieve properimpedance matching and reduced signal reflections along the feednetwork, it is necessary to construct transmission lines (and antennaelements) having intrinsic impedances tuned to 2-10Ω, and to maintainnear uniformity among the physical dimensions of all conducting elementsfound in the line, pad, and via structures. As described below, the useof round, small diameter discrete wires is desirable in constructing afrequency-selective antenna element. The characteristic impedance,Z_(o), of a wire over a ground plane is calculated per the formula:

Z _(o)=SQRT(L/C),  (1)

where L=2×10⁻⁷·ln(2B/R) and C=1.17× 10⁻¹⁷·∈_(R)/L, and B is the heightof the wire's center above ground and R is the wire's radius. Insertinga high-κ ceramic base dielectric layer 131 having a relativepermittivity ∈_(R)≧10, preferably ≧25, and even more preferably∈_(R)≧100, between the wire (antenna elements 117 and 123) and theground plane 133 makes the construction of a low impedance (2-10Ω)discrete wire antenna element from a small diameter (18 μm) wirepractical. (Similar techniques may be employed in constructing theinterconnect structure.) Lower impedance (2-5Ω), smaller round wirediameter (18 μm) feed networks require higher permittivity ceramiclayers 131 (e.g., ∈R=100 and 150), to achieve layer thicknesses of t=1.4and 3.3 μm. respectively, for 50 input impedance. These thicknesses arewithin reasonable existing manufacturing capabilities. Whereas, evenhigher permittivity ceramic layers (e.g., ∈_(R)=450) are able to achievean input impedance of 2Ω with a layer thickness on the order of 1 μm forcircuits constructed with discrete wire having a diameter ≦25 μm. Higherimpedance (10Ω) and larger diameter wire (>25 m), or the use of ribbonwire conductors, alleviates the need for a dielectric layer 131 withrelative permittivity ∈_(R)≧100 separating the wire (antenna elements117 and 123) from the ground plane 133, so high-κ dielectrics with∈_(R)≧60 are preferred for the high-κ dielectric layer 131 when largediameter (≧25 micron) wire or ribbon wire discrete wire conductors areincorporated into the antenna modules. In instances when the module isassembled from discrete components it is advantageous to use anyunderfill or adhesive agent 130 known to practitioners skilled in theart that mechanically secures the assembly into a solid structure.

The antenna elements may be disposed upon or embedded (completely orpartially) in composite dielectric body 129, which is composed of atleast one ceramic dielectric inclusion embedded in a low permittivitydielectric host material (∈_(R)≦10, preferably ∈_(R)≦4.5) comprised ofan organic dielectric, amorphous silica, pure alumina, titania,tantalates, and/or admixtures thereof. The dielectric inclusions have arelative dielectric permittivity ≧10 and/or a relative dielectricpermeability ≠1, that cause the effective relative permittivity of thecomposite dielectric body to increase without cause the reflectance ofthe body to become too strong. As shown below, the higher effectivepermittivity of the composite body ((∈_(REff)≧10, preferably ∈_(R)≧4)causes the physical lengths of any resonant antenna elements embeddedwithin it to be reduced by more than 50% and can also impart spectralfiltering to the antenna module. Various configurations of the compositedielectric body and conducting elements are possible. In one embodiment,the composite dielectric body simply comprises a base dielectric layerhaving a surface at which the electrically conductive element(s) is/aredisposed, and a dielectric host layer disposed above the base dielectriclayer and covering the at least one electrically conductive element. Insuch an embodiment, the dielectric inclusions may be embedded in eitheror both of the base dielectric layer and the dielectric host layer. Inother embodiments the composite dielectric body includes multipledielectric layers and optionally spacer layers therebetween. Theconductive elements and dielectric inclusions may be embedded as desiredin any of the additional layers.

FIGS. 2C-E provide greater detail on the electrical interconnectstructure used to establish electrical communication between the poweramplifier (or other) semiconductor die 135 and the antenna element(s).As referenced above, the interconnect structure may be comprised of aconventional interconnect structure that has an impedance-matchingnetwork (as well as other components) integrated within it. In apreferred embodiment of the invention (see FIG. 2C), the electricalinterconnect structure 137 contains a network of transmission lines 132positioned at a uniform distance above one or more metallization layersincluding ground planes 134 located at the top 134A or bottom 134B ofthe electrical interconnect structure 137 and embedded betweendielectric layers 120A,B. Electrical contact is established betweendevices located on the opposing major surfaces of the interconnectstructure 137 by conductive means 122 inserted into the dielectriclayers that attach the transmission line network 132 to contact pads 124located in, but electrically isolated from, the metallization layer thatforms the ground planes 134. The transmission line network 132 mayoptionally include pad structures (not shown) similar to 121A/B thatmight be alternatively used as terminations for transmission linestructures, capacitive elements, or localized stop layers to provide ameans to drill through a dielectric layer and insert conductive means122 to make electrical contact with a conductive element (transmissionline, antenna element or contact pad) situated on a dielectric surfaceplaced in a position superior to the transmission line network 132. Forimpedance matching reasons described below, it is preferable that thetransmission line network 132 have physical dimensions (diameter andcross-sectional form) similar to the antenna element, when that antennaelement is formed from a discrete wire. In embodiments comprising smalldiameter (≦50 μm) discrete wires, dielectric layers 120A,B composed ofhigh-κ ceramic dielectric are required to provide an electricalinterconnect structure with intrinsic impedance that is matched to thesource impedance of the semiconductor die. High-κ dielectric materialstypically have high loss factors. Therefore, it is an additionalpreferred embodiment of the invention to insert a lower loss dielectriclayer 126 between the dielectric layers 120A,B and the transmission linenetworks 132A,B associated with each dielectric layer as shown in FIG.2D. This lower loss dielectric layer 126 may comprise a pure organic orceramic dielectric material, or be a composite dielectric bodycomprising a pure organic or ceramic dielectric host material thatcontains dielectric inclusions that cause the effective relativepermittivity of the lower loss dielectric layer 126 to increase or matchthe relative permittivity of the base dielectric. In this configuration,conductive means 122 is used to make electrical connection between therespective transmission line networks 132A,B and the contact pads 124.Alternatively, the base dielectric layers could also be composed ofcomposite dielectric materials comprising a pure organic or ceramicdielectric host material that contains dielectric inclusions that causethe effective dielectric permittivity of the layers to become elevated.The present invention encompasses more complex interconnect structuresthat might require multiple interconnect structures 137 to be assembledone on top of one another to complete the circuit that is insertedbetween the semiconductor die 135 and the antenna element(s). The moduleis completed by affixing the at least one electrical interconnectstructure 137 to at least one semiconductor die 135 mounted on a thermalreservoir device 139 and an external circuit 140 using additionalconductive means 138, preferably through the use of a suitable underfillor adhesive agent 130. In order to minimize the module's footprint, itis preferable to position those contact pads used to make electricalconnection to the external circuit 124A as a peripheral array, whilelocating those contact pads used to make electrical connection to thesemiconductor device 124B to an interior region of the bottom groundplane 134B of the electrical interconnect structure 137. (See FIG. 2E).These principals may be alternatively applied to discrete wire ribbonsor patch antenna elements without the need for extreme permittivity(∈_(R)≧150) as the lower self-inductance of these structures naturallyleads to lower intrinsic impedance.

Although FIGS. 2A-2B depict a dual-band antenna module, wherein one ofthe dipole antennas 117, 123 might be used for transmit frequencies andthe other used for receive frequencies or, alternatively, wherein theyboth might be used to manage transmission (or reception) over a singlefrequency band, dual frequency bands, or multiple frequency bands of awireless interface, as is the case in 802.11g applications; i.e., thescope of the invention is not limited to single band, dual band ormultiple band antenna modules, or folded dipoles. An antenna module inaccordance with the invention might comprise a single straight dipole,an array of dipoles, a log-periodic dipole array, spiral antenna(s),trapezoidal, tear drop, or multiple dipole antennas folded in shapesthat are different from those depicted in FIG. 2A. Unless otherwisestated, it should be assumed that a particular metallization layer,whether used to form an antenna element, a transmission line, acapacitor or contact pad, or a ground plane structure is formed using athin film deposition technique, such as sputtering or evaporativedeposition, or electroplating, or a lower cost thin/thick filmdeposition screen printing or ink-jetting technique that locally appliesthe metallization using a conductive paste or ink. The specific designof an individual antenna element or array would be appropriatelydetermined by technical criteria for a specific application. The antennamodule's construction and architecture, and methods used to narrow thebandwidth of an individual antenna element, provide sufficient spectralfiltering to impart a high degree of frequency selectivity. Methodsprovided here allow miniaturization of overall dimensions, and improveperformance efficiencies by minimizing power loss in comparison toconventional RF front-end assemblies. Additional options includeshielding for the semiconductor die and individual antenna elements fromspurious electromagnetic signals emanating from each other as well asambient, or adjacent sources.

In general, the overall length of each dipole arm 117A, 117B is chosensuch that the combined length of the two arms is resonant at anoutput/input frequency of interest. For a dipole in free-space, theresonant length, l, varies as:

1˜λ/2,  (2)

where λ is the free-space wavelength of the center frequency ofinterest. It is well known to practitioners skilled in the art ofantenna design, that adding a capacitive load to the antenna will alsoshorten the resonant length, l, while increasing the antenna element'sbandwidth. This could be achieved by adding the capacitive load as aparallel plate capacitor (not shown) formed between the ground layer 133and the pads 121A, 121B terminating the dipole arms 117A, 117B. Sincevery thin layers of high-κ ceramic material separates the pads andground layer, substantial capacitive loads could be added using verysmall surface areas, providing significant latitude to enhanceminiaturization (by shortening the dipole arm length) and trim theantenna bandwidth.

It is well known to practitioners skilled in the art that dielectricloading will, for an infinite dielectric body, decrease the resonantlength as:

1˜J(2√∈_(R)),  (3)

where ∈_(R) is the relative permittivity of the dielectric body. Thus,if an antenna element is embedded within an infinite dielectric bodythat has an ∈_(R)=100, the resonant length of the dipole is reduced by afactor of 10 compared to its free-space resonant length. The antennaelement's bandwidth will decrease with increased dielectric loading, anda narrower bandwidth may also be achieved by constructing the antennaelement with smaller diameter wire. One technique provided by theinvention combines a wire bonding process, which positions discretewires with small diameters (e.g., current standards can utilize 18 μm)very precisely, with a manufacturing process that produces ceramicshaving dielectric properties not known to occur in nature. Existingtrace geometries used in prior art multilayer ceramic modules haveminimum dimensions on the order of 2 mil (50 μm), which limits frequencyselectivity, and imperfect surface roughness (≈5 μm RMS) andmetallurgical purity, which contribute to phase noise and resistive lossat higher frequencies. In contrast, wire bond conductors provide nearlyperfect micron scale geometric controls, negligible surface roughnessand the highest metallurgical purity possible, characteristics useful inovercoming many of the high frequency limitations of conducting elementsconstructed from metallic pastes. These properties permit constructionof antenna elements that are both frequency-selective (narrow bandwidth)and small in size, thereby eliminating the need for costly and lossgenerating filtering and switching components in an RF front-end, andreducing the footprint in mobile platforms, like cellular phones andlaptop computers, where means that reduce occupied volume have greatvalue. Numerical methods are used to determine precise specificationspertaining to the selection of the number of antenna elements, theirwire diameters, the dielectric properties and volume of high-κdielectric material used to surround the antenna elements, and thecapacitive loading that is used to terminate the discrete wire antennaelement to achieve certain size, bandwidth, and frequency performancetargets.

It should also be noted that the terminating capacitive pads 121A, 121Bshould be designed to accommodate manufacturing tolerances to achievereproducible control over the antenna element's center frequency.Mechanical deformation to wires used to form the antenna element(s)during the pay-out of a wire bond assembly can introduce slack in thelength of the wire between stitch bonding steps that are used to bondthe antenna element (discussed below). Minor variations in the wire'soverall length can significantly alter the antenna's tuning. Forinstance, wire slack in an antenna embedded in a dielectric material 129with an ∈_(R)=100 that induces the combined length of both dipoles armsto vary by ±10 μm will cause the resonant (center) frequency, to vary by±1 MHz. Therefore, wire bond machines with systems to manage bond lengthare preferred assembly tools. Furthermore, the geometry and size of thecapacitive pads should be designed not only to add capacitive load thattunes bandwidth and center frequency, but also to accommodatemanufacturing tolerance for wire bond length by becoming an integralcomponent that determines the dipole's resonant length. While there maybe circumstances wherein the terminating capacitive pads are shaped assimple rectangles while still accommodating manufacturing tolerancesaffecting wire bond length, in a preferred embodiment it is desirable touse a specific region of the pad to accommodate potential variations inwire bond length that will not affect the capacitor pad's feed point,nor substantially alter capacitive loading on the pad. FIG. 3 shows apossible configuration for the terminating pad 139 that minimizes anycapacitive load added to the pad structure to compensate for variationsin wire bond length without substantially affecting the pad's electricalfeed point. Pad 139 includes a narrow neck or “landing strip” 141 havingwidth W that is slightly larger (approximately 5 μm or less) than thediameter of bonded wire 143 and is sufficiently long to accommodate themost probable wire length 145A, 145B deviations from the desired thewire bond termination point 147. The length of the “landing strip” issufficiently long to provide a distance d that allows the point 148 atwhich the “landing strip’ adjoins the main pad structure 149 becomes theeffective feed point to the capacitive pad, regardless of whether thewire terminates at point 145A or point 145B. The landing strip ispreferably placed in electrical communication with perpendicular padareas 151A, 151B that comprise at least 95% of the capacitive load. Theperpendicular pad areas 151A, 151B need not be square or rectangular asshown in FIG. 3, but could assume triangular, digital, circular or anyarbitrary geometric form that provides sufficient capacitive load withinspace limitations provided by specified design criteria. As aconsequence of this configuration, the wire, landing strip, andcapacitor pad now determine resonant length l_(α) of the dipole arm.

In one embodiment, low permittivity materials like pure amorphous silica(α-SiO₂) or organic media (such as FR4, Rogers Duroid™, GTEK, or PFTETeflon™) having ∈_(R)=3.9 and 2.2≦∈_(R)≦4.6, respectively, can be usedas the host dielectric body for electrically conducting elements used toform the antenna or transmission line structures as well as ceramicdielectric inclusions that act as meta-material modifiers. In thisinstance, it is preferable to select the dielectric inclusion from agroup of materials having ∈_(R)≧9.3, which can include pure alumina,titania (TiO₂), tantalum oxides, niobium oxide, neodymium oxide,zirconium oxide, hafnium oxide, or any admixture thereof. In otherembodiments, a higher permittivity ceramic host (60≦∈_(R)≦100) havingmoderately low loss, such as pure alumina, tantalum pentoxide (Ta₂O₅,∈_(R)≅60) or titania (∈_(R)=90) might also be used as the hostdielectric. In this instance, the dielectric inclusions would comprisehigh permittivity ceramics (∈_(R)≧200), preferably, but not exclusively,selected from the group consisting of barium-strontium titanate ceramic(Ba_(X)Sr_(1-X)TiO₃), barium-strontium zirconate-titanate(Ba_(X)Sr_(1-X)Zr_(Y)Ti_(1-Y)O₃), and lead-lanthanum zirconate-titanate(Pb_(X)La_(1-X)Zr_(Y)Ti_(1-Y)O₃), where 0≦X≦1 and 0≦Y≦1. The dielectricinclusions acting as meta-material modifiers in either organic orceramic dielectric hosts may also comprise magnetic materials (μ_(R)≠1),such as ferrite (Fe₂O₃), alkaline earth ferrites (MFe₁₂O₁₉), where M isselected from the group consisting of Mg, Ca, Sr, Ba, or lanthanummanganates (LaMnO₃) doped with any element from the periodic table.

Organic host materials are attractive because of their lower cost, butcan often contribute higher loss and have dielectric parameters that arenot as stable as a function of frequency or temperature, so the use ofceramic materials is preferred. The host dielectric body could also becomposed of an admixture of amorphous silica (α-SiO₂) and titania(TiO₂), having a chemical composition Si_(1-x)Ti_(x)O₂, where 0≦x≦1, oran admixture of amorphous silica and alumina [1-X]•SiO2•X•Al2O3, where0≦X≦1. The ceramic host dielectric body, layered structures, or ceramicdielectric inclusion can be fabricated using liquid aerosol sprayprocesses, as disclosed in the de Rochemont patents discussed above,wherein the oxide ceramic is deposited at process temperatures rangingbetween 250° C.≦T_(sub)<475° C. using liquid aerosol sprays and areactive gas atmosphere consisting of CO/CO₂ in an inert carrier gas.This method is preferred because it allows a limitless variety of veryattractive ceramic compositions and metal components to be included intothe finished product at atmospheric pressures and high manufacturingthroughput rates. However, liquid aerosol sprays are not a suitabledeposition technique when organic polymers are used as the dielectrichost material. Organic dielectric hosts, which can be applied using aspray, require the use of low-temperature powder aerosol spray processesto form the ceramic dielectric inclusions. These powder aerosol sprayprocesses, well known to practitioners skilled in the art of ceramicdeposition, may consist of fine ceramic powders aerosolized in gas-jetspray that are deposited on a substrate held at temperatures as low asambient (room) temperatures under a vacuum atmosphere. Powder aerosolspray processes can also be used to fabricate the dielectric hostmaterial as well. Both spray techniques can be applied to form modulesin accordance with the present invention. However, the use of liquidaerosol metalorganic precursors is preferred for superior ability tocontrol microstructure (grain size) properties of the deposited ceramicas discussed below.

Amorphous silica is among the most transparent electromagnetic materialsavailable in nature. It has the lowest real dielectric permittivity(∈_(R)=3.9) among the environmentally safe ceramic oxides, and it has aroom temperature dielectric loss tangent tan δ=2×10⁻⁵ that is roughlytwo orders of magnitude better than the ceramics used in tape castassemblies. Titania is also a loss (tan δ=9×10⁻⁴) high-κ (∈_(R)=90)dielectric. However, both materials have such highly refractory thermalproperties that their use is prohibited in their pure form inconventional multilayer assemblies, as the temperatures neededconsolidate disparate powders incorporated into green tapes woulddestroy any high conductivity metallic elements designed into thestructure. Conventional silica-based or titania-based tapes must includechemical additives that lower the sintering temperature needed toconsolidate a tape stack to temperatures below the melting point of themetals incorporated into the assembly. These additives increase both thedielectric loss tangent (tan δ) and the real permittivity of adielectric body. The ability to fabricate amorphous silica attemperatures ≦450° C. provides a means to incorporate this ultra-lowloss material as a host material in a meta-material dielectric body,rather than as a supplemental layer, to increase the radiationefficiency of a miniaturized antenna embedded within the meta-materialdielectric.

FIG. 4 depicts a meta-material dielectric body 153 formed byincorporating one or more secondary phase material regions 155 within apure amorphous silica host 157. Secondary phase material regions 155preferably comprise ferrites, ferroelectric (comprising titanate,zirconate, tantalate, niobate) oxide ceramics, among others. Themeta-material dielectric body 153 will assume an average dielectricresponse that is a weighted sum of the fractional percentage of eachmaterial region incorporated in the dielectric body for frequencieswhere the physical dimension d of the secondary phase material regions155 is less than about 1/20th of the free space wavelength (λ/20) ofsuch frequencies, regardless of whether or not the secondary phasematerial regions 155 are randomly distributed or organized in a periodicarray. Electromagnetic radiation will diffusely scatter from themeta-material dielectric body 153 when the secondary phase materialregions 155 are randomly distributed throughout the dielectric host 157and have a physical dimension d that is roughly equivalent to theradiation's free space wavelength, i.e., >λ/10. Conditions for Braggscattering will be satisfied and an electromagnetic bandgap (EGB)material will be formed when these secondary phase material regions 155have physical dimensions d that are roughly equivalent (˜>λ/10) to thefree space wavelength and are periodically arranged with an appropriatespacing. Table I demonstrates the factor reduction in dielectric loss,leading to improved radiation efficiency, which is achieved whenamorphous silica and/or titania are used as the dielectric host, ascompared to typical tape cast ceramic hosts in a meta-materialdielectric. The values reflected in Table I are for meta-material bodiesformulated to contain 23% high-κ BST ceramic (∈_(R)=453) and aredesigned to have roughly the same effective real permittivity (˜100),however, the amorphous silica host provides 50% or greater lower lossover the typical tape cast ceramic host. A similar improvement occurswhen lower fractional volumes (5%) of higher permittivity, high losslead lanthanum zirconate titanate (PLZT, ∈_(R)=2000) is incorporatedinto the respective hosts. The inclusion of titania into the systemsprovides even more substantial improvements in real permittivity ∈_(R)with mild increases in dielectric loss tan δ.

TABLE 1 Effective Pure Dielectric Dielectric Properties PropertiesDielectric Material ε_(R) tan∂ εR_(Eff) tan∂_(Eff) Amorphous Silica(α-SiO₂) 3.9 2 × 10⁻⁵ — — Titania (TiO₂) 90 9 × 10⁻⁴ — —Polyfluorotetraethylene (PFTE) 2.1 4 × 10⁻⁴ — — Rogers Duroid (Duroid)2.3 5 × 10⁻⁴ — — LTCC Green Tape (TAPE) ≈5 2 × 10⁻³ — — Barium StrontiumTitanate 423 9 × 10⁻³ — — (BST) Lead Lanthanum Zirconate 2000 5 × 10⁻² —— Titanate (PLZT) α-SiO₂ (58%)/TiO₂ (42%) — — 40 4.3 × 10⁻⁴ PFTE(58%)/TiO₂ (42%) — — 39 6.1 × 10⁻⁴ Duroid (58%)/TiO₂ (42%) — — 39 6.7 ×10⁻⁴ TAPE (58%)/TiO₂ (42%) — — 41 1.5 × 10⁻³ TiO₂ (54%)/α-SiO₂ (46%) — —50 5.0 × 10⁻⁴ TiO₂ (54%)/TAPE (46%) — — 51 1.4 × 10⁻³ α-SiO₂ (77%)/BST(23%) — — 100 2.0 × 10⁻³ TiO₂ (77%)/BST (23%) — — 167 3.1 × 10⁻³ PFTE(77%)/BST (23%) — — 99 2.4 × 10⁻³ Duroid (77%)/BST (23%) — — 99 2.5 ×10⁻³ TAPE (77%)/BST (23%) — — 102 3.6 × 10⁻³ α-SiO₂ (95%)/PLZT (5%) — —100 2.3 × 10⁻³ PFTE (95%)/PLZT (5%) — — 99 2.9 × 10⁻³ Duroid (95%)/PLZT(5%) — — 99 3.0 × 10⁻³ TiO₂ (95%)/PLZT (5%) — — 181 3.3 × 10⁻³ TAPE(95%)/PLZT (5%) — — 102 4.4 × 10⁻³

Similarly, when H-field considerations are important to the antennadesign, it is advantageous for the meta-material to be composed ofdielectric inclusions that comprise materials (described above) thathave a relative permeability μ_(R)≠1.

For ease of understanding, reference is now made to FIGS. 5A-H toillustrate construction steps that may be used to create antenna modulesin accordance with the invention. A peel-apart metal foil 159, comprisedof a carrier foil 161, an adhesive barrier layer (usually a chromatefilm) 163, and an electroplated conductive metal film 165, is rigidlyattached to a flat, thermally conductive base 167. (See FIG. 5A). Foilsprovide a means to incorporate metallization layers that have extremelysmooth surface textures (≦0.5 μm RMS) that are not achievable whenforming modules from ceramic green tape processes. Photolithographytechniques are used to pattern contact pads 169 and ground planestructures 171 into the electroplated metal film 165 as need be to meeta specific design objective, as shown in FIG. 5B. A base dielectriclayer 173 with thickness t (shown in FIG. 5C) is then deposited on thesurface of the module using an aerosol spray process. The basedielectric layer 173 may comprise an organic dielectric, or a ceramicdielectric selected from the group consisting of amorphous silica, purealumina, titania, tantalum oxide and any admixture thereof, oralternately may consist of a composite dielectric body comprised of oneor more ceramic dielectric inclusions embedded in a an organicdielectric, or a ceramic dielectric selected from the group consistingof amorphous silica, pure alumina, titania, tantalum oxide, or anadmixture thereof. Preferred liquid aerosol spray processes aredescribed in U.S. Patent Nos. by de Rochemont et al. in U.S. Pat. Nos.6,027,826, 6,323,549, 5,707,715, 6,143,432 and 6,742,249, the contentsof which are all incorporated by reference in their entirety. As shownin FIG. 5D, the base dielectric layer 173 may comprise an organicdielectric, amorphous silica, titania, a high-κ ceramic oxide, or ameta-material dielectric body further comprised of ceramic oxidedielectric inclusions 177 embedded in a host dielectric material 179.The choice of dielectric used to form the base dielectric layer 173, andits thickness t, is determined by electrical design requirements neededto match the impedance of electrically conducting elements that will bedeposited on the surface of layer 173 with the source impedance of thesemiconductor die to which the electrically conducting elements willeventually be electrically connected. In accordance with the presentinvention, the meta-material dielectric body comprising b dielectriclayer 173 may be prepared by selectively spray depositing the dielectricinclusions 177, and subsequently spray depositing the organicdielectric, amorphous silica, pure alumina, or titania host material179, in contrast to prior art methods that backfill secondary dielectricmaterials that are drilled into host dielectric materials. Ink-jet spraytechnologies are the preferred methodology to selectively deposit thedielectric inclusions 177, but other methods, such as spraying through amask or screen, or spray a uniform layer that is selectively etchedusing photolithographic techniques can also be employed. Often, thesprayed dielectric material will have a rough or undulating surface,which can be flattened by applying one or more spin-coated layers 181 ofthe same or similar ceramic compositions when they are formulated fromaerosolized metalorganic precursor solutions, in order to fill in voidsor troughs and provide a surface finish that has a mean dielectric layersurface roughness of ≦±0.2 μm RMS, preferably ≦±0.1 μm RMS. A sprayedlayer is required to form a covalent bond and strong adhesion with themetal foil 159. Spin-coated metalorganic liquid films generally do notadhere well to metal substrates after thermal decomposition, but do bondwell when thermally decomposed on dielectric material that has the sameor similar composition. Spray methods are generally preferred to buildup thick deposits at significantly faster rates, whereas thermaldecomposition of spin-coated films is applied typically when it isdesirable to bring the oxide surface within a flatness or roughnesstolerance. Thus, a layer of dielectric material may be spun coat uponthe metallization layer to obtain, for example, a surface with asmoothness ≦1 μm RMS.

With reference to FIG. 5E, a variety of techniques such as, for example,thin film deposition, screen printing, etc., may then be used to affixterminating capacitor pads 183 to the oxide ceramic dielectric layer 173(which may or may not include the smoothing layer 181.) Vias 185 may belaser drilled to expose the contact pads 169 that have beenphotolithographically patterned into the electroplated metal film layer165. The transparency of amorphous silica facilitates machine indexingand circuit registration, thereby permitting minimization of thephysical dimensions of the contact pads to a major axis size on theorder of 50 μm. The reduced size contact pads provide a significantadvantage in reducing signal reflections at GHz frequencies. Multilayerceramic interconnects will typically require pad sizes on the order of8-10 mil (200-250 μm). Optically transparent amorphous silica dielectricmay be selectively applied to regions 182 in the immediate vicinity ofthe contact pads 169 (as shown in FIG. 5D) when forming the oxideceramic dielectric layer 173, in order to facilitate the use of opticalindexing techniques when forming the laser drilled vias. Immediatevicinity is understood to mean the area surrounding the contact padextending a distance that is sufficient to resolve the pad fully fromneighboring pad or ground plane structures upon optical inspection.

With reference to FIG. 5F, a microstrip transmission line structure(antenna or interconnect) can be formed by inserting a stud 187 can beinserted into each via 185 to make an electrical connection with acorresponding contact pad 169. The studs are roughly equivalent inphysical size to the contact pad 169 and discrete wire conductingelement 191, and respectively have heights that are roughly the same asthe thickness t of the oxide ceramic dielectric layer 173. In caseswhere the thickness t of the oxide ceramic dielectric layer 173 is lessthan about 5 microns, a discrete wire can be stitch bonded directly toeach exposed contact pad 169. Alternative means, such as back fillingthe vias with conductive paste or electroplated, or thin film metaldeposits, can also be used and may be preferred in applications thatrequire high I/O counts. Studs are preferred conductive means, however,in low I/O count module applications such as antenna where cost-valuebenefits are justified. In antenna module embodiment that comprisediscrete wire conducting elements, polymeric bonding posts 189 aretemporarily and selectively affixed to the surface of the basedielectric layer 173. These polymeric bonding posts allow precise stitchbonding of the discrete wire conductive elements 191 to the module. Eachconductive element 191 is bonded between one of the studs 187 and aterminating capacitor pad 183. The polymeric bonding posts 189 are usedto control sway in the discrete wire as it is drawn between the stud andtermination pad, and typically are placed every 10 mil (254 microns)along a straight line path, or at inflection points where it planned tointroduce a bend in the discrete wire conductor element 191. Eachpolymeric bonding post 189 is composed of low-temperature thermosetplastics that is sufficiently soft to allow the discrete wire conductorelement 191 to sag into it when a bonding tool 193 is drawn and broughtdown to a touchdown position proximate the surface of the module along adesired wire path such that the spooled wire traverses the polymericbonding post 189. The polymeric bonding posts 189 should, however,remain tacky enough to hold the wire in place as the bonding tool 193 isdrawn to the next touchdown position. In physical embodiments thatcomprise ceramic base or host dielectric layers, polymer formulationsfor the low-temperature thermoset plastic may vary and will depend onthe temperature used to stitch bond the discrete wire elements, whichcan range from 150-250° C., however many commercially availableformulations are widely available. In these embodiments, thelow-temperature thermoset plastic preferably contains additives thatpromote cross-linking after a UV-cure step that results in rigidfixation of the discrete wire conductor element 191 even at temperaturesas high as 350° C. experienced in subsequent processing. Many suchadditives are well known to practitioners skilled in the art of polymerformulation.

Reference is made to FIGS. 5G&H for embodiments that do not containorganic dielectrics. In this instance, once the discrete wire conductorelements have been stitched, the module is heated to a temperaturebetween about 300° C. and 350° C. to in a processing step of selectivedeposition of ceramic bonding posts 195 using liquid precursors thatpermanently affix the discrete wire conductor elements 191. (See FIG.G). These ceramic bonding posts can also be applied at room temperaturesusing aerosolized powders. As shown in FIG. 5H, the polymeric bondingposts 189 are subsequently removed by thermal decomposition, and themodule is then heated to temperatures ≦450° C. An additional dielectriclayer 197 may be optionally sprayed on top of the module, as shown inFIG. 5I. The additional dielectric layer 197 may be composed of organicdielectric, amorphous silica, pure alumina, titania dielectric, or ameta-material dielectric including an organic dielectric, amorphoussilica, alumina, or titania host comprising dielectric inclusions, andthe layer has a thickness judiciously selected to satisfy designobjectives. FIG. 5J illustrates a fully assembled antenna module 199that contains a dielectric inclusions selectively located in preferredregions of the module. In embodiments that do not include organicdielectric and are formed from aerosolized liquid precursors, the spraydeposited ceramics will have an amorphous microstructure. Theseembodiments may then be heated to temperatures ranging between 500 and900° for a period of about 15 to 60 minutes in a redox atmosphere inorder to sinter the high-κ ceramic elements into a desired crystallinephase and real permittivity value, ∈_(R). An advantage to using anamorphous silica or an titania host is that it is negligibly affected bythis sintering step. Crystallization may also be applied by localizedheating using laser, or focused beams from infrared or ultraviolet lightsources. In embodiments that contain an organic dielectric the ceramicdielectric inclusions are applied using a powder aerosol spray, whereincrystallization and microstructure control (grain size) is imparted tothe ceramic dielectric inclusion in its powdered state prior to theaerosol spray deposition step and there is no need to apply subsequentthermal processing steps post aerosol spray deposition. The carriermetal foil 161 may then be separated from the flat, thermally conductivebase 167, and then removed with the barrier layer 163 from the ceramicantenna module 199. As shown, the contact pads 169 and ground planestructures 171 remain embedded in the base dielectric layer 173 afterthe carrier metal foil 161 and barrier layer 163 are removed from themodule. Similar methods are used to build the interconnect structureused as the electrical interconnect structure 137 described in FIG. 2 byadding the further steps of drilling additional vias through anyadditional dielectric layers 120B or 126 and 120B to expose atermination pad or transmission line network 132, inserting conductivemeans 122 into the vias exposing said transmission line network ortermination pad. In single strip-line embodiments shown in FIG. 2C, ametallization layer is disposed on dielectric layer 120B to provide aground plane 134A and contact pads 124. In double strip-line embodimentsshown in FIG. 2D, the metallization layer is disposed on dielectriclayer 126 to form a second transmission line network 132B that is inelectrical communication with the first transmission line network 132Asituated between base dielectric layer 120A and dielectric layer 126 byconductive means 122. In this instance, a second base dielectric layer120B is then deposited on top of the second transmission line network132B. Holes may be drilled through said second dielectric layer 120Binto which additional conductive means 122 is applied to form a via thatprovides electrical communication with contact pads in metallizationlayer 134A that is also used to provide a second ground plane.Reproducing these steps to provide additional structures similar toelectrical interconnect structure 137 on top of the existing one, as thecase may be depending upon circuit complexity is a straightforwardextrapolation and should be considered as falling under the scope of thepresent invention.

FIGS. 6A&B describe more generic embodiments, wherein the antenna module192 may consist of one or more conducting antenna elements 194, whichmay comprise a thin or thick film or at least one discrete wireterminated on a capacitor pad, embedded in a composite ceramicdielectric body 196 situated on the surface of a metallization layer 198that contains portions which act as a ground plane 200 and otherportions that serve as contact pads 201 in electrical communication withthe conducting antenna element 194 through a conductive means 202. Themetallization layer 198 may alternatively, as discussed below, beintegral to the surface or body of any additional electronic component,such as a semiconductor die or interconnect that acts as an electricalinterconnect structure, or be fabricated as a detachable artifact asshown in FIG. 5, that is later assembled into a more complex module asshown in FIG. 2. The composite ceramic dielectric body 196 may comprisea single base dielectric layer 203 on which the conducting antennaelements 194 are positioned. It may alternatively comprise the basedielectric layer 203 and antenna elements 194 with an additional spacerdielectric layer 204 applied on top of both. Additionally, the antennamodule 192 may comprise the base dielectric layer 203, spacer dielectric204 and an additional dielectric layer 205, or the base dielectric layer203 and the additional dielectric layer 205 without the spacerdielectric layer 204, as shown in FIG. 6B. The base dielectric layer 203and the additional dielectric layer 205 can be selected from the groupconsisting of organic dielectric, or a ceramic dielectric selected fromthe group consisting of amorphous silica, pure alumina, titania,tantalum oxide and any admixture thereof, or alternately may consist ofa composite dielectric body comprised of one or more ceramic dielectricinclusions embedded in a an organic dielectric, or a ceramic dielectricselected from the group consisting of amorphous silica, pure alumina,titania, tantalum oxide, or an admixture thereof. It is preferred thatspacer dielectric layer 204 comprise organic dielectric, or a ceramicdielectric selected from the group consisting of amorphous silica, purealumina, titania, tantalum oxide and any admixture thereof. As shown inFIG. 6B, the base dielectric layer may also be used to electricallyisolate regions of the metallization layer 198 by situating it inregions 206 located between the ground plane regions 200 and contact padregions 201. FIG. 6B also shows embodiments that comprise a monopoleantenna where a single contact pad is used to establish electricalcommunication with the conductive antenna element 194. FIG. 6C shows amore specific embodiment of the invention wherein the base dielectriclayer 204 comprises a composite dielectric body wherein opticallytransparent amorphous silica ceramic is located in the vicinity 207 of aground plane region 200 where it is adjacent to a contact pad region 201to improve registration while inserting a conductive means 202 toestablish electrical communication between said contact paid and theconductive antenna element 194. FIG. 6C also makes reference to theapplication of spin-coated ceramic dielectric layers used to reducesurface roughness to levels less than or equal to ±0.2 μm RMS at theinterface between a metallization and a ceramic layer 208 or theinterface between one dielectric ceramic layer and another dielectricceramic layer 209.

The present invention provides additional means for improving theradiation efficiency of an antenna by reducing reflective loss atdielectric interfaces incorporated into the antenna module.Electromagnetic waves traversing the interface between one dielectricregion and another will have a portion of their energy reflected at theboundary in accordance with Snell's Law. Internal reflection becomesquite strong at dielectric boundaries that have large differences inrelative permittivity, thus causing electromagnetic transmission acrossthe interface to be sharply reduced. For instance, if the antennaelement were embedded in a conventional (non-meta-material) high-κ(∈_(R)≅100) dielectric body, the difference in dielectric densitybetween the high-κ material and air (∈_(R)≅1) would cause 96% of theelectromagnetic power to be internally reflected at the interface andonly 4% of the energy to be transmitted across the boundary producing anunacceptably low radiation efficiency for the structure. The use of lowpermittivity dielectric hosts, such as organic media or amorphous silicawhich have ∈_(R)≦4.5, provide a “clear window” that improves the powertransmitted into the composite ceramic dielectric body. Dielectricinclusions can be added in sufficient number and volume to produce aneffective relative permittivity (∈_(REif)) that will alter the antennaelements resonant properties. Amorphous silica is preferred as the hostmaterial for the composite dielectric body because it simultaneouslyprovides low permittivity (∈_(R)=3.9) and ultra-low loss (tan ∂=2×10⁻⁵).A meta-material composite dielectric comprised of an amorphous silicahost that contains finer scale repetitions of high-κ ceramic dielectricinclusions sufficient in sufficient quantity to produce an effectiverelative permittivity of ∈_(R)≅100 forms a dielectric interface with airthat allows 89% of the incident electromagnetic power to transmitted.11% of the power is internally reflected back into the dielectric mediumfrom which it originated. Strong internal reflections will occur at theinterfaces between the amorphous silica host and the high-κ dielectricinclusions, but, as these bodies are small compared to the radiationwavelength, the reflections principally affect wavefront phasing in themanner described above when discussing the defect resonator. Maximaltransmission efficiencies occur at a dielectric interface where thedifference in the relative permittivity values between the lowerdielectric density and the higher dielectric density materials isreduced. Therefore, as shown in FIG. 7A, a further embodiment of theinvention comprises an antenna module 192 containing at least one low-κcap layer 210, having a relative permittivity less than the module'scomposite ceramic dielectric body 196, or any host dielectric that iscontained therein, that is applied on radiating surface 211 of thecomposite ceramic dielectric body 196 that improves the transmissionefficiency of electromagnetic emissions emanating from a conductiveantenna element 194. For example, if the low-κ antireflective cap layer210 comprises polyfluoro-tetraethylene (PFTE-Teflon), which has arelative permittivity (∈_(R)≅2.2), and the composite meta-materialdielectric body contains amorphous silica (∈_(R)=3.9) as its hostmaterial, then 96% of the power incident upon the PFTE-Air interfacewill be transmitted across this first dielectric boundary, and 98% ofthe power incident upon the PFTE-Silica interface will be transmittedacross this second dielectric boundary. Thus, the total powertransmitted across the combined structure is 94% (98%×96%). Adding aplurality of cap lower permittivity layers 212 (FIG. 7B) to thestructure to produce a step wise or continuous gradient in permittivityto further reduce internal reflections across the radiating surfacefurther improves electromagnetic transmission efficiencies. Dependingupon design objectives and the host dielectric material contained in thecomposite dielectric body 196, these lower permittivity layers maycomprise organic, ceramic, or a combination of organic and ceramicmaterials. Specific lower permittivity layers useful in embodiments thatcomprise an amorphous silica dielectric host within the compositedielectric body 196 would include, but are not limited to, or more ofthe following materials polyfluoro-tetraethylene (Teflon) (∈_(R)≅2.2),bis-benzocyclobutene (BCB) (∈_(R)≅2.7), polyvinyl formal (∈_(R)≅3),polyvinyl butyral (∈_(R)≅2.6), Rogers Duroid (∈_(R)≅2.3), and polyimideKapton (∈_(R)≅3.1).

With reference to FIGS. 8A-8D, the electrically conducting thin or thickfilm antenna elements applied to form a conducting antenna element 194may take the form of spirals 213, log-periodic arrays 215, tear drop216, or patches 217 (shown in circular form for example purposes only)with or without one or more voided or slotted regions of selectablegeometric shapes, such as squares 219 or triangles 221, or anycombination thereof, depending on the bandwidth specification of aparticular design objective.

Reference is once again made to FIG. 2 to illustrate methods to tune theimpedance of the antenna module 114 to match source or input impedanceof a power amplifier (PA) or low noise amplifier (LNA) (or other)semiconductor die 135, respectively, that may be integrated within themodule body. Often, the source or input impedance of these semiconductordie are in the range of 2-10Ω, so tuning the impedance of the antennamodule 115 to match semiconductor die 135 requires the conductingelement(s) 117,123, and 132 to be located very close to ground. This isparticularly true when the semiconductor's impedance falls in the rangeof 2-5 Ω and a round discrete wire having a very narrow diameter (˜18-25μm or less) is used as a conductive element in the circuit. In thisinstance, the base dielectric layers 120A/B and 131 should comprise ahigh-κ dielectric layer (∈_(R)≧10, preferably ∈_(R)≧60) with a thicknesst on the order of 1-3 μm or more to separate the conducting elements117, 123, and 132 from any ground plane contained within metallizationlayers 133 or 134(A,B).

Often any gain that is realized in transmission line efficiency bymatching the antenna's load impedance to the semiconductor die's outputor source impedance is negated by the inefficiency of a radiatingelement when it is located that close to an electrically conductingground. As is well known to practitioners skilled in the art of antennadesign, an antenna element located over a perfect electricallyconducting (PEC) ground plane will radiate roughly 50% of its poweroutwardly away from the ground plane, and roughly 50% of its power willbe directed towards the ground plane, where it is reflected back towardsthe antenna element. The electrical component of the reflectedelectromagnetic wave undergoes a 180° phase shift as it is bounced offthe ground plane. It is preferred to locate the radiating element aquarter wavelength (λ/4) above the PEC ground plane so that the phase ofthe reflected electromagnetic waves is phase delayed an additional 180°in phase and will, thus, constructively interfere with the signalsemanating from the radiating element. Consequently, electromagneticsignals that are reflected from a PEC or electrically conducting groundplane that is located very close to the radiating elements willdestructively interfere with signals emanating from the antenna in realtime, causing the radiator to have zero or negligibly low radiationefficiency as nearly 50% of the radiated power destructively interfereswith the 50% of the power that is reflected off of the ground plane.

This situation is reversed, when the radiating elements are located overa ground plane constructed to act as an artificial magnetic conductor(AMC). AMC's are alternatively described in the art as high impedancesurfaces, Sievenpiper surfaces, or perfect magnetic conductors (PMC).With reference to FIGS. 9A&B, an AMC 223 is assembled by disposing adielectric body 225 over an electrical ground plane 227, applying anarray of capacitor pads 229 to the dielectric surface opposing theelectrical ground plane 227, and inserting electrically conductive vias231 between the array of capacitor pads 229 and electrical ground thatplace them in electrical communication with the electrical ground plane227, wherein the array of capacitor pads 229 are centered on saidelectrically conductive vias 231. The capacitor pads 229 need not besquare as shown, and can possess arbitrary shape or form that has amaximal physical dimension I<P such that a gap of dimension g existsbetween said capacitor pads Resonant characteristics of the AMC aredetermined by the specific properties of the dielectric body 225 and theconfiguration of the capacitor pad array. Under resonant conditions, theE-field components of electromagnetic (EM) waves incident upon the AMCsurface are reflected from the ground plane without a shift in phase. Asa result, placing the antenna closer to the AMC ground plane enhancesconditions for constructive interference between those reflected EMcomponents with the E-field components radiated by the antenna indirections away from the ground plane surface. The incorporation of AMCstructures into the antenna module's ground plane, therefore, allows theintrinsic impedance of radiating antenna elements to be reduced tolevels that more closely match the impedance of PA/LNA (or othersemiconductor) die. This embodiment of the present invention enables theconstruction of a transmit/receive (T/R) module with far greater overallsystem efficiency compared to conventional systems, as it eliminates theneed for impedance matching networks and additional loss generatingcomponents.

The characteristic frequency response of AMC surface to EM wavesincident upon it and propagating across the surface is depicted in FIGS.10 and 11, respectively. The reflection coefficient of EM waves incidentupon the AMC ground plane induces a phase shift in the reflectedelectric field component that varies from +180° to −180° over afrequency band, f₀ to f₁, producing a 0° phase shift at a centerfrequency f_(c). The radiation efficiency of an antenna elementconfigured parallel to and very near to the AMC ground plane ismaximized at f_(c). While it is preferable to configure the AMC groundplane such that f_(c) coincides with the antenna's resonant frequency,this is not always possible when multiple antennas operating at multiplefrequencies are configured over the same AMC ground plane. Therefore, itis generally advantageous to design the AMC ground plane such that theoperational frequencies of one or more antennas positioned over the AMCoverlap the range of frequencies wherein the AMC induces a ±90° phaseshift in reflected electric field components, with a ±450 phase shiftbeing preferred.

The AMC ground plane will also suppress the propagation of parallelelectric and magnetic components of EM waves across the AMC surface overa range of frequencies f₂ to f₃. This suppression of surface-wave modesthereby allows multiple antennas to operate independently of one another(without inductive interference) when configured over the same AMCsurface, whether or not the antennas operate at the same or nearlysimilar frequencies. The range of frequencies, f₂ to f₃, over whichradiation will not couple into adjacent components is an attribute ofthe AMC's design and does not necessarily related to or match the phaseshift frequency band. This range of frequencies, f₂ to f₃, isalternatively referred to as the AMC bandgap. Thus, the AMC ground planemay be designed such that the operational frequency band(s) of one ormore antennas configured upon it overlap(s) the AMC bandgap.

The design of the AMC ground plane 223 is influenced primarily by thesurface area of the ground plane and periodicity P of the capacitorpads, the gap g between the capacitor pads 229, the height h (length)and diameter of the vias 231, and the properties of the dielectricmedium 225 separating the capacitor pads 229 from the electricallyconducting ground plane 227. The AMC center frequency is generallydetermined by:

f _(c)=√(L/C),  (4)

where L is the inductance generated by the electrically conductive via231 short to electrically conducting ground plane 227 within anindividual capacitor pad 229, and C is the capacitance of the individualcapacitor pad. The AMC bandwidth is generally affected by the propertiesof the dielectric medium 225. In general, a broader AMC bandwidth isachieved when the dielectric medium 225 has low dielectric loss and lowreal permittivity. Amorphous silica provides the lowest realpermittivity (∈_(R)=3.9) and lowest dielectric loss (tan δ=2×10⁻⁵) ofall known ceramic dielectric materials. AMC bandwidth is also broadenedwhen high permeability materials (∈_(R)≠1) are incorporated into thestructure. The large size of resonant AMC structures imposes a practicallimitation against deploying them in mobile devices. As described above,high permittivity materials can be used to reduce the physicaldimensions of resonant structures. Therefore, methods and productsenabled by the present invention that combine high permittivity(∈_(R)≧9.3) ceramics, with high permeability (∈_(R)≠1) materials andamorphous silica to minimize an AMC's physical dimensions whilemaximizing the bandwidth are highly desirable. A specific objective ofthe invention is to construct a meta-material antenna module thatincorporates an AMC ground plane to improve impedance matching withsemiconductor die to which the antenna element is coupled. A furtherembodiment claims an AMC ground plan structure that comprises the highpermeability and high permittivity dielectric layers and/or inclusionsembedded within or disposed upon a dielectric host having a loss tangentless than 5×10⁻³, or what is generally possible with green tape ceramicstructures. Suitable hosts include organic dielectrics, such as PFTETeflon, Rogers Duroid, and Rogers GTek, which will typically have losstangents in the range of 1.5-3×10³. The use of an amorphous silica hostis a preferred embodiment of the invention as it has a loss tangent of2×10⁻⁵. Specific reference is now made to FIGS. 12A thru E to detailthis aspect of the invention, which describes an antenna module 228 thatcomprises at least one reduced impedance conducting antenna element 230situated directly upon a high-κ ceramic dielectric layer or inclusion232A and an AMC 233 acting as the ground plane for the module, whereinthe AMC 233 may comprise conventional dielectric materials or compositedielectric bodies enabled by this invention. In another aspect, theinvention describes an AMC 233 having a composite dielectric body 234that comprises a low-loss dielectric component, in whole or in part,having a loss tangent less than 5×10⁻³, preferably less than 1×10⁻³. Aspecific objective of the present invention is the construction of anAMC ground plane 233 comprised of a composite dielectric body havingdielectric components selected from the group consisting of amorphoussilica, organic dielectrics, high permittivity ceramic dielectricshaving ∈_(R)≧9.3, high permeability ceramic dielectrics having μ_(R)≠1,that has a physical size ≦80% and an operating bandwidth greater than orequal to a similarly configured AMC constructed without the use ofamorphous silica or high-κ or high-μ ceramic dielectrics. In addition tothe details described above, this embodiment might optionallyincorporate a composite dielectric body wherein the high-κ or high-μceramic dielectrics are incorporated as discrete inclusions within a lowloss dielectric host 235. Alternatively, the ceramic dielectrics may beincluded within the composite dielectric body 234 as a high permittivityceramic layer 232B or high permeability ceramic layer 236. The ceramicdielectric layers and/or inclusions may be located anywhere withincomposite dielectric body 234 that optimizes the resonantcharacteristics of the AMC to meet design specifications for aparticular application. The composite organic dielectric body 234 couldbe constructed by applying dielectric layers inclusions to apre-fabricated sheet of organic dielectric, drilling through holesthrough the structure and applying metallization using techniques wellknown to practitioners skilled in the art, such as plating techniques,among others, to form the capacitor pad array 223, vias 231, andelectrical ground plane 227. Similarly, composite ceramic AMC's mayalternatively be fabricated by starting with a substrate, such as quartzor any other suitable inorganic material listed above, wherein theadditional ceramic/metallization components are similarly applied usingpowdered or liquid aerosol techniques. Alternatively, dielectriccomponents of the AMC may be fabricated entirely from aerosol sprayprocesses deposited on a metallization layer, with optional patterningif needed, wherein the metallization layer itself may comprise adetachable layer, such as a peel-apart foil, or may be affixed to asurface of a component, such as an interconnect or semiconductor die,that remains integral to the finished antenna module. In cases where anorganic dielectric body is assembled using a peel-apart foil toformulate a ground plane 227 and contact pads 240 in electricalcommunication with vias 239 (if needed), the ceramic dielectric layersand/or inclusions are applied by a powdered aerosol spray techniques,and organic dielectrics are applied by liquid aerosol deposition,preferably. Vias are formed by drilling through the depositeddielectrics. Metallization is applied to the holes and to form the padarray using plating or other means. Metallization foils can also be usedto fabricate AMC structures containing ceramic host dielectrics.Although the ceramic structures can be fabricated entirely with liquidor powder aerosol techniques, the ability to prepare highly uniformdielectric host materials comprised of amorphous silica, titania, purealumina, tanatalum oxide, niobium oxides or admixtures thereof, andhigh-κ/high-μ ceramic dielectrics with controlled grainsize/microstructure is a principal advantage to using liquid aerosolspray techniques. An additional benefit to the use of liquid precursorsand all-ceramic structures is the optional ability to apply spin-coatedlayers to form extremely smooth surfaces 237, having roughness less thanor equal ±0.2 μm RMS at an interface between a sprayed ceramicdielectric and any additional material body (metal, dielectric, air)that might be disposed thereupon to improve physical characteristicsthat are influenced by interfacial quality. The antenna module, asdepicted in FIGS. 12A thru E, may optionally include an additionalcomposite ceramic dielectric body 238 disposed thereupon that maycomprise any one or all of the attributes depicted in FIGS. 6&7, and theconducting antenna element 230 may comprise a dipole antenna element,patch, or monopole antenna, as described herein. Electricalcommunication with the conducting antenna element 230 is managed withelectrical feeds that may optionally be configured as vias 239 embeddedwithin the AMC body 233 to provide electrical contact between at leastone contact pad 240 within metallization layer 230, wherein the via 239and the contact pad 240 are electrically isolated from the electricalground plane 227 and the array of capacitor pads 240. Alternatively, theelectrical feeds may comprise transmission lines 241 that traverse thesurface of high-κ 232A rather than penetrating the AMC body 233 to makeelectrical connection through at least on side 243 of the antenna moduleas shown in FIGS. 12 A&D. The via(s) 239, contact pad(s) 240, andtransmission lines 241 need be located as depicted in FIG. 12, butpositioned anywhere within the antenna module 228 that provides optimaldesign characteristics. These vias can be backfilled with conductivepastes, or, preferably loaded with a column of studs to build viastructures that have roughly the same physical dimensions as the antennaelement and the contact pads. There are, however, certain instances whenit is advantageous to use a discrete wire to maintain electricalcommunication with the at least one contact pad 240 to the conductingantenna element 230. For example, when the antenna element(s) 230consist of discrete wire, it is preferable to use a continuous discretewire to connect the contact pad 240 to the terminating capacitor 222. Asshown in FIG. 12E, a ball bond 244 is formed by inserting the wirebonding tool's capillary into the laser drilled via and bonding the wireto the at least one contact pad 240. In this instance, it is preferredto construct an AMC ground plane structure 233 that that has a height hgreater than 100 microns, preferably greater than 150 microns, so thatthe discrete wire may having flexible bending point 245 that allows thewire to be folded and stitched, as shown in FIGS. 5A-J, along the AMCsurface. This minimum height is preferred to avoid mechanical weaknessgenerated in the discrete wire as the ball is flamed off of the wire'send. The flame-off procedure causes mechanical weakness in a region 246identified by practitioners skilled in the art of wire bonding as a HeatAffected Zone (HAZ), which can extend 100-150 microns beyond the ball,depending upon flame-off conditions. There is a high incidence of wirefracture at kinks or bends that are applied to the HAZ part of the wire,so it is preferred to deposit wire for a length greater than the HAZbefore bending the wire to run parallel to the AMC (or other) groundplane structure. After completing the electrical contact between thecontact pads 240 and the discrete wire antenna element's termination pad222, the drilled vias in which the discrete wire and ball bond 244 aresubsequently filled with the dielectric host material 247.

It is well-known to practitioners skilled in the art of antenna designthat resonant antenna elements provide enhanced performance at multiplefrequencies where the reactive component of the antenna's impedance isclose to zero. For instance, the reactive component of a resonantdipole's input impedance will be zero-valued at resonant frequencies,defined as those frequencies wherein the combined length of the dipolearms, l, are sequenced multiples of the frequencies' half-wavelengthsgiven by:

1≈{λ_(R)(n+1)}/2,n=0,2,4,  (5)

As a consequence, an antenna module (configured either as 192 or 228)designed to operate exclusively at a primary resonant frequency may alsobe susceptible to higher order resonant frequencies. In the case of thedipole, the primary resonant frequency would have a wavelengthλ_(Ro)≈21, but will also be sensitive to signals at electromagneticfrequencies having wavelengths corresponding to λ_(Rn)≈21/(n+1), wheren=2, 4, 6, . . . , that can cause noise and interference when notproperly filtered. An antenna module deployed in a wirelesscommunications system is usually surrounded by additionalmicroelectronic components placed adjacent to the antenna module thatwill radiate spurious signals that have the potential to correspond toboth the primary and higher order resonant frequencies. A specificobjective of the present invention is the construction a reducedcomponent-count antenna module as shown in FIGS. 6 and 12 that does notrequire additional components to isolate the signals of primary interestto the wireless communications device. To address this need, thecomposite dielectric body 196 and 238 optionally added to embodiments192 and 228, respectively, may be engineered to act as a frequencyfilter that rejects the higher order resonant frequencies from near andfar-field sources, as well as spurious signals at resonant frequenciesfrom microelectronic components in the vicinity, but not placed directlyin front, of the module's conducting antenna element. A compositedielectric body 196 or 238 may be configured as an electromagnetic bandgap (EBG) meta-material dielectric to establish a means for suppressingsuch spurious signals. With reference to FIGS. 13&14, an EBGmeta-material dielectric 251 comprises a periodic array of high-κdielectric inclusions 253 within a lower-κ host dielectric host 255. TheEBG produces a stop band 257 to electromagnetic (EM) frequencies thathave a wavelength λ_(c) comparable to the spacing period P between thedielectric inclusions 253. The dielectric inclusions 253 containedwithin a low permittivity host dielectric 255 will stronglyscatter/reflect EM waves incident upon them. Electromagnetic wavepropagation is inhibited when the periodic spacing P between theinclusion causes the scattered components to destructively interferewith the transmitted EM wave components, those not incident upon theinclusions, over frequencies ranging between f_(L), the band's lowerstop frequency, and f_(U), the band's upper stop frequency. A stop bandwill typically be characterized by the frequency, f_(c), located at thecenter of the band. The width of the EGB stop band 257 varies as afunction of the ratio of the relative permittivity between thedielectric inclusions 253 ∈_(R(incl)) and the low-k dielectric host 255∈_(R(host)). Conditions for destructive interference over a broadfrequency band is generally maximized when the ratio of the relativepermittivity between dielectric inclusion and the dielectric host is inthe range 10≦∈_(R(incl))/∈_(R(host))≦50, and when low loss materials areused to form the inclusions as well as the host. The EBG meta-materialcomposite dielectric 251 is fabricated by applying an optional base hostdielectric layer 255A to a substrate 248 using an aerosol spray.

The substrate 248 may be an independent dielectric or conducting layer,or such a layer that is part of an antenna module, semiconductor die,microelectronic devices, etc., or it may be a sacrificial layer thatallows the EBG meta-material composite dielectric 251 to be separatedfrom the substrate 248 as a free-standing body. The present inventionuses an organic dielectric, preferably a low-loss organic like RogersDuroid, GTek, or PFTE Teflon, or a ceramic dielectric, preferably aceramic dielectric having loss less than 5×10⁻³, as the host materialfor the EGB meta-material composite dielectric body 196/238. Ininstances, where large size reductions of the antenna module (comparedto its free-space equivalent) is desired, titania, which is reported tohave a relative permittivity ∈_(R)=90 and a loss tangent of 9×10⁻⁴,tantalum oxide, pure alumina, or any admixture thereof with amorphoussilica, is preferred as a dielectric host medium. Dielectric inclusions253 are selectively deposited on the optional base host dielectric layer255A or the substrate 248. Titania (TiO₃) is preferred as the dielectricinclusion 253 material in low permittivity host dielectrics, likeorganic media or amorphous silica. The dielectric inclusions 253 canhave any shape, ceramic composition, and physical dimension thatmaximize stop band performance for a specific design objective. Thecenter frequency f_(c) of an electromagnetic wave having wavelengthλ_(c) is used to fix the band location. In general, the period P shouldnot be greater than 0.15λ_(c) or less than 0.05λ_(c), and preferably,not greater than 0.12λ_(c) or less than 0.08λ_(c). The high-κ dielectricinclusions 253 may assume any geometric form, rectangular, circular,etc., and comprise a single ceramic composition, giving them all uniformdielectric properties, or multiple ceramic compositions, wherein theinclusions' dielectric properties are varied in accordance with theirrelative position within the inclusion array. It is generally preferredto define the inclusions 253 by a major dimension a and a minordimension b, where the major axis a is generally not greater than0.075λ_(c) or less than 0.0257λ_(c), preferably not greater than0.06λ_(c) and less than 0.04λ_(c), and the minor dimension b isgenerally in the range of 0.5 a to 0.01 a, preferably 0.25 a to 0.075 a.The various parameters P, a, b, and ∈_(R(incl)) are selected bynumerical analysis to suppress the interfering signals anticipated fromspurious emissions by neighboring components to penetrate the module andshield the antenna elements. In embodiments that incorporate an organicdielectric host, the inclusions 253 are formed using aerosol powderspray deposition. Selective location can be applied using a variety ofmethods, such as spraying through a mask, or spraying a uniform layerand selectively removing portions of the sprayed layer usingphotolithographic techniques to reveal the appropriate inclusion formand dimension. Aerosolized powder deposition techniques may also be usedto form inclusions in all-ceramic embodiments, however, liquid aerosoldeposition methods are preferred for its superior microstructure controland faster processing speeds. Ceramics sprayed from liquid aerosols mayalso be applied through masks or with photolithographic processing,however, the use of ink-jet spray deposition methods is preferred toimprove manufacturing efficiencies and when inclusion complexity(multiple dielectric compositions) is a design objective. Laser trimmingmethods are applied to ensure inclusion size and shape meet designtolerances. The EBG meta-material dielectric 251 is completed byapplying dielectric host materials over the assembly using liquid orpowder aerosol techniques. In all-ceramic embodiments, spin-coatedlayers having surface roughness ≦±0.2 μm RMS may be optionally appliedat interfaces 247 and 249 to improve design tolerances.

Reference is now made to FIGS. 15&14, which depicts an antenna module258, additionally comprising at least one semiconductor die 259A and aheat sink 260, in electrical communication with an interconnect circuit261 and additional semiconductor die 259B,C, wherein at least one EBGmeta-material dielectric body 262 is situated in the vicinity of theantenna module 258 to isolate spurious electromagnetic signals 252emanating from neighboring semiconductor die 259A,B or discrete EMsources 254 that are not located in the antenna module's radiation beamprofile 263. The EBG meta-material dielectric body 262 may also belocated as a dielectric shielding layer 262A that is part of anyinterconnect circuit 261 to which the antenna module is connected orsituated near, or as a shielding material affixed to a container (notshown) in which these circuits are housed. The EBG meta-materialdielectric body will completely null the electromagnetic emissions atfrequencies falling within the stop band. This provides a superioralternative to conventional shielding methods which can introduce systemand phase noise through ground loop perturbations. An EBG meta-materialdielectric body 262 so used should be engineered to have a stop band 257with a lower stop frequency f_(t). that is less than the antenna'sfundamental (n=0) resonant pass band, and extends to an upper stopfrequency f_(U) that is greater than pass band(s) one of thehigher-order (n=2, 4, 6, . . . ) resonant frequencies f_(R2), f_(R4),f_(R6), . . . , etc.

Making reference to FIGS. 16, 17, & 18, an additional embodiment of theinvention claims an antenna module 264 that comprises a conductingantenna element 265 separated from a metallization layer 266, whichcontains portions that function as a ground plane 267 and contact pads268 in electrical contact through conductive means 269 with theconducting antenna element 265, by a base dielectric layer 270. Thisembodiment further comprises a composite dielectric body 271, consistingof a dielectric host 272 and a plurality of dielectric inclusions 273having relative permittivity greater than the dielectric host 272,wherein the composite dielectric body 271 is configured to act as an EBGdefect resonator and the conducting antenna element 265 is locatedwithin an inclusion-free zone 273 of the composite dielectric body 271.An EBG resonator comprises a standard EBG 251 (see FIG. 13) with atleast one dielectric inclusion 253 removed from the periodic arraycontained within the host dielectric 255. The defect resonatorsubstantially limits propagation of electrical fields to a narrow passband 277 of frequencies around the center frequency f_(c) of the stopband that would exist if the EGB meta-material did not contain adefective “inclusion-free zone”. In effect, the inclusion-free zonecauses the EBG stop band to split into a Lower Stop Band 279 (betweenf_(Llower) and f_(Lupper)) and an Upper Stop Band 281 (betweenf_(Ulower) and f_(Uupper)). The EM field components of the pass bandfrequencies are also strongly localized to regions within theinclusion-free zone, and will typically be attenuated by 5-20 dB withina first perimeter 283 extending beyond the first nearest row ofdielectric inclusions 273 (roughly a distance of one (1) period P fromthe center of the inclusion-free zone), and be attenuated by >40 dBbeyond a second perimeter 285 extending beyond the second nearest row ofdielectric inclusions 273 (roughly a distance two (2) periods P from thecenter of the inclusion-free zone). The field localization and the widthof the narrow pass band 277 is a complex function of the ratio of thereal permittivity ∈_(R(incl))/∈_(R(host)) of the high-κ dielectricinclusions 273 and the dielectric host 275. Superior localization andfrequency filtering is achieved when the dielectrics used for form thecomposite dielectric body 264 have reduced loss tangent. In thisembodiment of the invention, the conducting antenna element 265 issituated in the center of the inclusion-free zone 273 and the compositedielectric body serves a dual purpose: in one aspect it is used as ahigh effective permittivity dielectric that miniaturizes the antennaelements size while providing high radiation coupling efficiency, inanother aspect it shields the antenna element from undesirable signalnoise and interference. In this instance, it is advantageous to tune theresonant conducting antenna element 265 to have its fundamentalresonance frequency, f_(R0), to be near (≦±2%) the center frequencyf_(c) of the EBG defect resonator. Often, it is not possible to designan antenna to have a sufficiently narrow band profile that roll-offs assharply with frequency as it is de-tuned from resonance to meet specificdesign objectives. A specific embodiment of the invention comprises anantenna module 264, wherein the composite dielectric body configured asa defect resonator 271 is designed to have a narrow pass band 277 thatis narrower than the antenna element's resonant band profile 285.Preferred embodiments of the invention utilize an EBG defect resonatorstructure that uses low loss (loss tangent ≦5×10⁻³) materials in thedielectric host 273 and the high-κ dielectric inclusions 273. Theseembodiments would comprise Rogers' Duroid, GTek or PFTE Teflon organicdielectrics, and low permittivity amorphous silica ceramic as hostmaterials, and low-loss titania-based (TiO₂) inclusion dielectrics.These titania inclusion compositions could alternatively by modified toobtain optimal relative permittivity ratios, ∈_(R(incl))/∈_(R(host)),with minor concentrations (0-5 mol %) of tantalum oxide (Ta₂O₅),zirconium oxide (ZrO₂), neodymium oxide (Nd₂O₅), hafnium oxide (HfO₂),and lead oxide (PbO) additives, by mixing said additives with titaniumoxide precursors in the formulation that is spray deposited to form aperiodic array of high-κ dielectric inclusions. Said additives are addedin amounts such that the ratio between the relative real permittivity ofsaid sprayed high-κ dielectric inclusions to the amorphous silica hostfalls within the range 25≦∈_(R(incl))/∈_(R(host))≦40.

It is often desirable to maximize the efficiency of the module forselected EM field polarizations. FIG. 19 shows a dipole antenna havingarms 301A, 301B and their associated feed points 303A,303B co-locatedwithin the inclusion-free zone 305 that is centered within an EBGmeta-material dielectric defect resonator 307. The arms 301A, 301B ofthe dipole are straight and may be oriented such that the majordimension a of the high-κ dielectric inclusions 309 are parallel withthe major dimension of the arms 301A, 301B. In another configurationshown in FIG. 20. the antenna may be centered within an inclusion-freezone 311 of an EBG meta-material defect resonator 313 as a folded dipole315, such that folded antenna arms 316A,316B have conductive segments ofnearly equal lengths oriented in the x and y direction and, therefore,the antenna is receptive to circularly polarized, or cross-polarizedelectromagnetic waves. In this instance, high-κ dielectric inclusions317 may be embedded within the reduced permittivity host dielectric 319in the shape of crossed rods (+), having two major axes that are alignedto be parallel with the major dimensions of the folded dipole arms316A,316B. Alternatively, a spiral or circular antenna element orelements 321 (see FIG. 21) may be configured within or on top of theinclusion-free zone 323 of the EBG meta-material defect resonator 325,and cylindrical symmetry may be imparted to the shape of the high-κdielectric inclusions 327 consistent with the electrical field patternsgenerated by the circular or spiral antenna element(s), which similarlyhave their feed points at and through the center of said inclusion-freezone. In each embodiment incorporating the antenna element at the centerof the inclusion-free zone of an EBG meta-material defect resonator, itis a preferable to provide a meta-material dielectric array thatcontains at least two rows of high-κ dielectric inclusions around saidinclusion-free zone(s).

Four non-limiting approaches will now be described by which a wirelesscircuit module 115 comprising at least one PA/LNA (or other)semiconductor die 135 may be connected through an interconnectingelectrical interconnect structure to the antenna element 114 (see FIG.2B) in accordance with the present invention. With reference to FIG. 22,semiconductor die 329 may be flip-chip mounted through conductive means331 directly to the antenna module 333, wherein the contact pads 328 areembedded in the surface of a base dielectric layer 327 to allowdielectric material 332 to be situated between metallization materialused to form contact pad 328 and ground plane 336 structures, such thatcontact pads 328 and ground planes 336 do not protrude from surfaces 339of the antenna module. An adhesive underfill 330 is used to secure thedie's placement to the module and to protect metal contacts fromenvironmental corrosion. This configuration is only suitable when thedie's signal inputs/outputs (I/Os) are designed and engineered to matewith the antenna I/O's without requiring a separate interconnectingelectrical interconnect structure 137 as shown in FIG. 2B or when aninterconnecting electrical interconnect structure is integrated into thesemiconductor die 329/135. However, this is rarely the case, and thereis often the need to interpose an interconnecting interconnect structure137 between the die 135 and the antenna module 114 to complete thewireless circuit module 115. Although the present invention describesnovel embodiments and methods to provide an electrical interconnectstructure 137 several interconnecting interposer interconnect structurestructures composed of organic or ceramic dielectric currently availablein the marketplace and could be used and fall within the scope of thepresent invention. Conventional interconnect structures contain contactpads 101 (see FIG. 1.), that protrude from the dielectric surface 100and require the use of solder masks 107 to effectuate flip-chipmounting. The use of the solder mask 107 often places an upper limit onthe pad density that can reliably applied when the die is flip-chipmounted. Semiconductor devices with high pad densities will often bebackside mounted to the interconnect network, and wire bonds will beused to establish electrical connection between the high-density pads onthe die and the electrical interconnect circuit. Wireless circuits arefrequently used in mobile platforms, where it is advantageous tominimize the size/footprint of the circuit. Therefore, it is desirableto develop a means to eliminate the need to apply a solder mask to allowhigh-density semiconductor die to be flip-chip mounted to wirelesscircuit module. In addition, currently available interconnect structureswill also contain pad structures having physical dimensions that aresubstantially different from the conductor traces and vias to which theyare electrically conducted, thereby causing undesirable signalreflections and losses internal to the interconnect at higher signalfrequencies.

Reference is now made to FIGS. 2B-D & 23A-E, to illustrate an objectiveof the present invention that provides a wireless circuit module 115comprising an antenna module 114, an interconnecting electricalinterconnect structure 137, with contact pads 124 that are embedded inthe surface of a base dielectric layer 120 to allow dielectric material341 to be situated between metallization material used to form contactpad and ground plane structures, such that contact pads and groundplanes do not protrude from surfaces of the interconnect structure orantenna module. It is a further objective of the invention that contactpads 124/328 have dimensions roughly equivalent to the size of thetransmission lines to which they maintain electrical communication.FIGS. 23A-E, depict the methods used to construct an interconnectingelectrical interconnect structure 342 as a separate body by depositing abase dielectric layer 343 on the surface of a peel-apart metal foil 345that is affixed to a substrate tool 350 that provides thermal controland mechanical stability to the assembly constructed thereupon. Groundplane structures 347 and internal contact pads 349A used to connect thesemiconductor die with electrical interconnect structure and externalcontact pads 349B used to connect the system to the electricalinterconnect structure 342 are been patterned into the electroplatedfilm of the peel-apart foil. The base dielectric may comprise amorphoussilica, pure alumina, titantia, a tantalum oxide, and an admixturethereof, or it may comprise a meta-material dielectric that incorporatesamorphous silica or an organic polymer as the host dielectric. Toconstruct low impedance (2-5Ω) transmission lines using very narrowdiameter wire, the base dielectric layer should comprise a thin layer ofhigh-κ ceramic oxide (such as layer 173 as shown in FIG. 6). This basedielectric layer may also comprise a meta-material dielectric body thatwith the requisite effective permittivity, however, a base dielectriclayer consisting of a high-κ ceramic with uniform dielectric propertiesis preferred. Holes 351A,351B are drilled into the base dielectric layer343 to expose contact pads 349A,349B, to which conductive means,preferably studs, 353A,353B are attached. Conductive inks or pastes areapplied to affix metallic termination pads 355A used to make verticalelectrical contacts. Termination pads 355B may be optionally appliednear the conductive means 353B to insure the quality of electricalcommunication with external connections 349B. Methods similar to thosedescribed above to place discrete wire conducting elements may beutilized in constructing discrete wire transmission lines 357A,357B,except that the capacitor pads described above that terminate theconducting wire would here would also comprise termination pads355A,355B that serve as transmission line termination points. It ispreferred that the termination pads 355A,355B have roughly the samedimension as the transmission lines 357A,357B, and, optionally, thestuds 353A,353B. As shown in FIG. 23B, it is also possible to useconventional trace structures 358A,358B fabricated from conductive inksor pastes, that may optionally including inductor coils, to electricallyconnect backfilled vias 352A,352B and termination pads 355A. Theinterconnecting interconnect structure is completed by depositing asecond base dielectric layer 359 and, optionally, a patterned groundplane structure 361 using screen printing techniques over the embeddedconductor structure. Wells 363A,363B may be drilled through the secondbase dielectric layer 359 to expose the termination pads 355A,355B. Thevertical connection to the top surface of the interconnectinginterconnect structure can be made by filling the wells 363A,363B with aconductive material, though it is preferable to deposit studs 365A,365Binto the wells in order to minimize the physical dimension of the viastructure. The interconnecting interconnect structure is completed byremoving the carrier layer and stop layer of the peel-apart foil fromthe assembly, resulting in the structure shown in FIG. 23C. The assemblyof an interconnecting interconnect structure that comprises anadditional low loss dielectric layer 126 as shown in FIG. 2D is astraightforward extension from the base structure 354 assembled in FIGS.23A&B. Making reference to FIGS. 23D&E, a low loss dielectric layer 356from the group consisting of amorphous silica, pure alumina, titania,tantalum oxide and any admixture thereof, or a meta-material dielectriccomprising an organic, amorphous silica, or pure alumina hostdielectric, is applied to the base structure 354. It is a preferredembodiment to spray deposit regions of optically transparent amorphoussilica 360A,360B in the vicinity of termination pads 355A used to makevertical interconnections so that automated optical imaging (AOI)techniques can be used to employ high precision laser drillingtechniques to construct vias 362 filled with conductive means that mayeither be a metal plating, a stack of coined studs (as depicted in FIG.23D), or backfilled conductive pastes or inks. The interconnectingelectrical interconnect structure 340 is completed by applyingtransmission lines 357A/357B, optionally applying termination pads 355Ato make additional vertical interconnections, and a second basedielectric layer 359. As shown in FIG. 23E, the interconnectingelectrical interconnect structure 340 is completed by drilling holes 364through the second base dielectric layer 359, inserting a conductivemeans 366, which may comprise a stud, plated metal, or backfilledconductive paste or ink, a top metallization layer 368 that is used toform ground plane regions 370 an contact pad regions 372, and separatingthe assembly structure from the sacrificial bodies 342 used as athermal/mechanical substrate to support the build.

In certain instances, it desirable to eliminate inductive coupling andcrosstalk, over specific frequency bands, between transmission linesassembled within the interconnecting electrical interconnect structure.Making reference to FIGS. 24&23A-E. the methods used to assemble an AMCground plane described above (see FIG. 12A) are employed to construct abase AMC ground plane 374, upon which an interconnecting electricalinterconnect structure 376 is assembled. Substituting the base AMCground plane 374 for the sacrificial layers 342 described above providesa means to suppress surface modes that cause inductive coupling betweenadjacent components, such as transmission lines, situated upon the AMCsurface 378. The base AMC ground plane thus remains integral to the bodyof the reduced-crosstalk interconnecting electrical interconnectstructure 380.

With reference to FIGS. 25A-D, wire bonding methods may be employed toproduce an interconnecting interconnect structure that electricallyconnects but mechanically decouples the PA (or other semiconductor) diefrom the antenna (or other) circuit. Wires 367A,367B may be attached bymeans of stitch bonds 369A,369B to terminating pads 371A,371B at thebottom of wells 373A,373B and controlled lengths of wire may be spooledso as to protrude from the top surface 375 of the second base dielectriclayer 377. Balls 379A,379B may be formed at the end of the wire(s) usingflame-off techniques. The wire structures depicted are sometimesreferred to as “inverted ball bonds” or “teed golf balls.” The ball andwire of the inverted ball bonds are then wick-coated with organicmaterial coatings 381A,381B of a type known to those skilled in the artas an addition polymer, which have the property of remaining intactuntil it is heated to a pyrolytic temperature, at which point theaddition polymer decomposes cleanly, leaving no measurable residue onthe coated surface. The interconnecting interconnect structure iscompleted by applying a liquid coating 383 of metalorganic precursors toa ceramic oxide to the top surface 375 of second base dielectric layer377 in sufficient quantities to fill the wells 373A,373B and to producea liquid coating that rises to at least 25% of the height of the ball inthe inverted ball bond. (See FIG. 25B,C). The addition polymer coatings381A,381B are selected to have a pyrolytic temperature that exceeds 250°C., in order to permit thermal decomposition of the liquid metalorganicprecursor coating 383 into a solid ceramic oxide layer 385 (as shown inFIG. 25C). A ground plane metallization layer 387 is selectivelyapplied. The addition polymer coatings 381A,381B are subsequentlyremoved from the inverted ball bonds by heating the structure to atemperature above 250° C., thereby exposing balls 379A,379B which aredisposed within mechanically protective wells 389A,389B, and theinterconnecting electrical interconnect structure is separated from thesacrificial layers 342. This interconnect structure configuration solvesa significant mechanical problem experienced in RF circuits. Frequently,PA (or other semiconductor) die generate a great deal of heat (>10 W).Silicon-based die have low coefficients of thermal expansion(CTE=2.6×10⁻⁶° C.⁻¹). Considerable shear stress is induced in RFamplifier die when they are coupled to conventional tape castinterconnecting interconnect structures, which typically havecoefficients of thermal expansion on the order of CTE===8-10×10⁻⁶° C.⁻¹.Therefore, the heat generating amplifier (or other semiconductor) diemay be coupled to an interconnecting interconnect structure formed fromamorphous silica (CTE≈1×10⁻⁶° C.⁻¹). However, as illustrated in FIG.25D, in instances where it is necessary to electrically connect a PA (orother semiconductor) die 391 to circuit element 393 with which there isa strong CTE mismatch between the semiconductor die 391 and the othercircuit element 393, but there is a good CTE match between the circuitelement 393 and the interposing connector, an interconnecting electricalinterconnect structure 395 may be included that contains inverted ballbonds 379A,379B that provide a mechanically decoupled electricalconnection between the CTE-mismatched elements, as the inverted ballbonds can mechanically flex within the wells 389A,389B that remain afterthe decomposition of said addition polymer organic material.

FIG. 26 depicts an alternative embodiment to the wireless circuit module115 in which the methods described in the present invention to assemblean antenna module (see FIGS. 5A-J, FIGS. 12A-E, and FIGS. 6A-C) andthose methods described to form an interconnecting electricalinterconnect structure (see FIGS. 23A&E and FIG. 24) are consolidated tofabricate an antenna module 386 that comprises an elemental ceramicantenna portion 388 and an interconnecting electrical interconnectstructure 390 integral to the module's body. In this embodiment, thewireless circuit module 392 consists of one (or more) semiconductor die394 in electrical communication with the antenna module 386.

Reference is now made to FIGS. 27A&B, which depicts a wireless circuitmodule 396 that, by virtue of the low deposition temperatures (<450° C.)of ceramics formed with aerosol sprays, is monolithically integrated atthe wafer scale onto a singulated semiconductor die 397. The wirelesscircuit comprises an integrated circuit 398 embedded into thesemiconductor wafer 399, an electrical interconnect structure 400, andan antenna module 401. Circuit cells defined by reference lines 402 aredefined to identify the physical boundaries of an individual integratedcircuit 398 and mark where the semiconductor wafer 399 will be scribedto form singulated die 397. The electrical interconnect structure 400and antenna module 401 portions wireless circuit module 396 areconstructed on top of each integrated circuit 398 using the methodsdescribed above prior to scribing the wafer semiconductor 399, intosingulated die 397. To provide for a direct off-chip electricalconnection to the wireless circuit module 396, conductive means 403 thatpenetrates the reference lines 402 may be optionally embedded into theelectrical interconnect structure 400 prior to die singulation so thatsaid conductive means 403 remains exposed on a surface 404 of thesingulated die. The low process temperatures enabled by the inventionprovides a means to prepare the fully integrated structure withoutsubstantially altering delicate dopant or materials structures that areimplanted, diffused, or deposited in or on the semiconductor material.Fully integrated structures may be assembled on silicon-based wafers(MOS or CMOS), silicon-germanium semiconductors, III-V compoundsemiconductors, and II-VI compound semiconductors, as well ascarbon-based semiconductors.

Many wireless systems will transmit and receive data over multiplefrequency bands. For instance, modern cellular phones designed tooperate in North America will typically, but not necessarily, managesome voice communications at the AMPS band, centered at 860 MHz, manageother voice and data communications at the PCS band, 1850-1930 MHz. Somecell phones will additionally provide global communications services atthe GSM bands, GSM-400, GSM-800 and GSM-1800, located in the 400, MHz,800 MHz, and 1800 MHz frequency bands, respectively. Other cell phonesmay provide some or all of the above communications protocols along withwireless internet access through 802.11a,b,g (2400 MHz and 5000 MHz) orglobal positioning systems at 2200 MHz). A given frequency band is oftendivided into transmit (Tx) and receive (Rx) bands, with AMPS Tx-modeoperating between 824-849 MHz and its Rx-mode operating between 869-894MHz. Conventional antenna assemblies have bandwidths that overlap two ormore of these communications bands, thereby requiring extensivefiltering and switching functions to be engineered into the RF front-endadding cost and complexity to the original equipment manufacturer (OEM).FIG. 28 is a representative description of an electrical block diagramthat is roughly consistent with the prior art used in multi-band RFfront-end that may typically be found in a cell phone or other wirelessdevice assembly. Such antenna assemblies 405 will often include, amongother components that are not specified in the figure, an antenna 407, afiltering stage 409 that includes one or more diplexers (among othercomponents) to separate frequencies of interest from those received bythe antenna, switches 411 that toggle the signal path from transmit toreceive modes, impedance matching networks signal isolators 413, lownoise amplifiers 414 (in the receive-side channel) and power amplifiers415 (in the transmit-side channel) that are electrically connected to asignal processing unit and/or transceiver controls 417. The fullcomplexity and cost of such circuitry grows as additional communicationsfrequencies and services are added to the assemblies.

Prior art RF front-end assemblies can often occupy a sizeable footprintand include large part counts that impose conversion costs. They canalso introduce signal loss that consumes appreciable power. Forinstance, SAW filters, which are commonly used to switch the radiofunction from Tx-mode to Rx-mode within a specific band, will typicallyimpose 3 dB of power loss. All of these attributes to conventionallyengineered front-ends are undesirable when engineering mobile platformslike cellular phones. Therefore, methods that reduce the size andconversion costs and power consumption and provide equivalent functionsas a conventional RF front-end are desirable in mobile platformapplications. In particular, methods that eliminate the need for a SAWfilter in the RF front-end extend the allotted power budget by 50% (3dB). Three non-limiting approaches are now described to furnish an RFfront-end module providing reduced component count, form factor, orpower loss enabled by the incorporation of at least onefrequency-selective antenna tuned to operate narrowly within a specificcommunications frequency band. FIG. 29A depicts the first suchembodiment, wherein a multi-band RF front-end module 406 eliminates theneed for a filter stage 409 that isolates different communications bandsof interest through the use of individual frequency-specific antennas408A, 408B, . . . , 408N tuned to each of the communications bandsdesired by the module's operational design. The individualfrequency-specific antennas 408A, 408B, . . . , 408N in turn feedswitches 410A, 410B, . . . , 410N that, (making reference to FIGS.29B,C), may optionally switch between the transmit 412A and receive 412Bchannels of, for instance, a cellular or Bluetooth communications band416, or to a single low-noise channel 418 that is used for both send andreceive as is the case in 802.11 (or similar) wireless communicationsband 420. The multi-band RF front-end module 406 may optionally includeimpedance-matching networks 413, if necessary, as well as low-noisepower amplifiers 414A, 414B, . . . , 414N on the receive-side, poweramplifiers 415A, 415B, . . . , 415N, on the transmit side, and(optionally) transceiver/signal processing die 417. FIG. 30 depicts analternative RF front-end module 419 configuration that exploits theimpedance matching and high-Q antenna characteristics enabled by theinvention wherein individual frequency-selective antennas are tuned tospecific transmit or receive frequencies, thereby eliminating the needfor filtering stages, isolators, and switching banks. This embodimentincludes a bank of power amplifiers 421A, 421B, . . . , 421N in directelectrical communication with a frequency-selective antenna 422A, 422B,. . . , 422N that has a bandwidth tuned to each transmit frequency, abank of low-noise amplifiers 423A, 423B, . . . , 423N in directelectrical communication with a frequency-selective antenna 424A, 424B,. . . , 424N that has a bandwidth tuned to each receive frequency, and,optionally, a signal processing/transceiver chipset 425,

An alternative, lower-complexity architecture for a multiple-frequencyRF Front End 427 useful in software radio applications is shown in FIG.31. In this instance, voltage-controlled oscillators 429A,B are used tomodulate the amplitude and phase inputs of a single power amplifier 431on the transmit-side, synthesizing modulated signals at the frequenciesof interest for signal transmission, f_(T1), f_(T2), . . . , f_(TN). Thesignal output of power amplifier 431 is in direct electrical contact toan array of frequency-selective antennas 433A, 433B, . . . , 433N,wherein each one of the frequency-selective antennas in the array istuned to have a narrow-band resonance at one of the frequencies ofinterest for signal transmission, f_(T1), f_(T2), . . . , f_(TN). Asimilar configuration can be implemented on the receive-side, whichcomprises an array of frequency-selective antennas 435A, 435B, . . . ,435N, wherein each one of said frequency-selective antennas is tuned tohave narrow-band resonance at one of the frequencies of interest forsignal reception, f_(R1), f_(R2), . . . , f_(RN), and may include anadditional frequency-selective antenna 437 that is tuned to resonate ata frequency of interest for global positioning systems (GPS),f_(RGPS)=2.5 GHz. The receiver array is separately configured to be indirect electrical connection with a low-noise amplifier 439 and to beelectrically isolated from the transmit-side of the wireless device. Theoutput of the low-noise amplifier 439 may optionally be directed througha signal mixing or filtering stage 441 to separate the individualfrequencies of interest for signal reception, f_(R1), f_(R2), . . . ,f_(RN), and f_(RGPS), before directing them to the central processingunit 443.

At present, the need to use acoustic wave filters in cellular phones toswitch between the transmit (“talk” or “send” mode) and the receive(“listen” mode) limits signal transmissions between the mobile platformand broadcast tower to only one direction at a given moment in time. Themobile platform is either in its listen mode or its talk mode. Thiscommunications protocol applies to analog voice as well as digital voiceand data streams, and is commonly known as half-duplex mode,Telecommunications service providers are finding increasing demand forcontent to be delivered to the mobile terminal. The limitations imposedby half-duplex mode communications restricts the quality of advancedcontent services, such as bidirectional audio visual streamingdeliverable to both ends of the communications link. As a result, videophone telecommunications often appear to be choppy or staggered. Theability to implement RF front-end architectures such as depicted inFIGS. 30 and 31 enables simultaneous full-duplex communications atmultiple frequencies without the need for extensive signal processingand embedded software, as high-Q frequency selective antennas may beconfigured to launch voice/data communications narrowly over thetransmit frequency channel without interfering with voice/datacommunications captured by the receive frequency channel. Thus, RFfront-end architectures as described herein will be extremely useful inwireless communication device services, such as, for example, fullduplex analog or digital voice/data streams or bidirectional audio videostreaming over one or multiple simultaneously operating wirelesscommunications frequency bands, which is a specific need for WCDMAand/or WEDGE 3G telecommunications protocols.

A fundamental problem in the practical application of high-Q circuits isthe need to maintain parametric stability. The precise value of therelative permittivity, ∈_(R), of high-κ dielectric body into which theantenna is embedded has a very strong effect on the frequency tuning ofthe antenna element. The center frequency f_(o) of a dipole antennaelement of length l varies roughly as:

f _(o) ∝c×1/(2√∈_(R)),  (6)

where c is the speed of light. For instance, if the medium into whichthe antenna element is embedded has an effective relative permittivityof ∈_(R)=100, than a 10% change in the effective relative permittivityto ∈_(R)=110 or 90, will cause f_(o) to shift by approximately 20%. A 1%change in the relative permittivity to ∈R=101 or 99 will cause f_(o) toshift by approximately 0.5%. Because the pass band is so narrow (1% orless), any fluctuations in manufacturing tolerances or thermal inducedchanges that would cause the value of the effective relativepermittivity ∈_(R) to change by 1% or more are undesirable because theywould cause the antenna's desired pass band tuning 445 to shift out ofthe desired pass band 447 as shown in FIG. 32.

Three fundamental parameters will cause changes to the value of theeffective relative permittivity ∈_(R) of a dielectric meta-material intowhich an antenna element might be embedded. First is the fractionalvolume of high-κ material incorporated into the meta-materialdielectric. The present invention utilizes deposition techniques thatcontrol the fractional volume of high-κ materials to volumetrictolerances within ±0.1%. Ink-jet spray methods may be used that have theability to control the location and sprayed volume of dielectricinclusions very precisely. Laser trimming techniques may also be appliedto ink-jetted dielectric inclusions to ensure volumetric tolerances areheld within ±0.1%, preferably ±0.05%.

The second parameter relates to compositional uniformity of the high-κmaterials incorporated into the dielectric meta-material. High-κmaterials that contain a single metal oxide component and a singlevalence state, such as tantalum pentoxide (Ta₂O₅), zirconia (ZrO), orhafnia (HFO), are preferred because they will exhibit no significantcomposition-dependent fluctuations in relative permittivity, ∈_(R), ifmetals purity levels to the precursors are held to values better than99.99%, or, preferred values of 99.999%. Other single component high-κmaterials known to have multiple valence states, such as, niobia (NbO,NbO₂, Nb₂O₅), or titania, (0, Ti₂O₃, TiO₂), are also preferred needsbecause they will exhibit no significant composition-dependedfluctuations in relative permittivity, ∈_(R), when redox processingconditions are sufficient to maintain uniform valence throughout theink-jetted deposit(s) and metals purity levels to the precursors areheld to values better than 99.99%, or, preferred values of 99.999%. Thecomparatively low relative permittivity of single component high-κmaterials, typical ∈_(R)≦90, requires relatively high volumetricfractions of these materials to be used to achieve meta-materialdielectrics having modest effective relative permittivity, which is notoptimal for many designs. Therefore, the use of multi-component high-κdielectric materials that have substantial relative permittivity,∈_(R)≧200 is preferred in many instances. Multi-component high-κdielectric materials would include, but are not limited to,barium-strontium titanate (BST) ceramic, with compositions spanning(Ba_(X)Sr_(1-X)TiO₃), barium-strontium zirconate-titanate (BSZT)ceramic, with compositions spanning (Ba_(X)Sr_(1-X)Zr_(Y)Ti_(1-Y)O₃), orlead-lanthanum zirconate-titanate (PLZT), with compositions spanning(Pb_(X)La_(1-X)Zr_(Y)Ti_(1-Y)O₃). A significant drawback to using theseceramic compositions is that they are susceptible to strong fluctuationsin compositional uniformity that can have dramatic effects on relativepermittivity at both the microscopic and macroscopic domain. This isespecially true for powder processed ceramics. Liquid aerosol spraytechniques, however, allow an unlimited number of precursors to bechemically mixed at the molecular level in solution. This degree ofchemical uniformity in the distribution of various precursors throughoutthe metalorganic solution is replicated in the spray-pyrolyzed deposit,enabling chemical composition of extremely high complexity to bedeposited with an, as yet, undeterminable high degree of chemicaluniformity.

The third parameter that impacts the temperature (and frequency)stability of the high-Q antenna structures disclosed under the presentinvention relates to the temperature dependence of the materialsincorporated into the meta-material dielectric. To be useful in cellularhandsets, it is necessary for the antenna module to hold its centerfrequency at operating temperatures ranging from −40° C. to 85° C. Toachieve this performance standard, the materials used to construct themeta-material dielectric should have a thermal dependence in theirdielectric constants that varies less 2×10⁻⁴° C.⁻¹ to achieve a ±1%variation in the relative permittivity at the temperature extremes, ifthe meta-material has an effective dielectric constant ∈_(R)=100 at roomtemperature (20° C.). This is particularly the case for the high-κmaterials. Thermal dependence in the expansion coefficient anddielectric constant of amorphous silica is within this tolerance overthe prescribed temperature range, which makes it an ideal host for themeta-material dielectric body. High-κ ceramics, particularly themulti-component compositions, are more problematic and highlight theneed for liquid aerosol spray deposition techniques. FIG. 32 [from Vest,Ferroelectrics, 102, 53-68 (1990)] correlates the thermal dependence inthe relative permittivity in BST ceramic prepared from liquidmetalorganic precursors. Immediately following metalorganicdecomposition high-κ ceramic prepared from liquid precursors has anamorphous structure with no crystallographic structure. Crystallinequality evolves with subsequent heat treatment. A very short sinteringcycle [≦10 minutes at 900° C.] produces ceramic with a very fine grainsizes (≅0.035 micron, 35 nm), see lowest curve 449 in FIG. 33. Thissmall grain size causes the relative permittivity to be suppressed,∈_(R)=: 200, as opposed to the more typical value of ∈_(R)≅400, which isfound when a longer cycle [20 min at 900° C.] evolves ceramic withmoderate grain size (≅0.10 micron, 100 nm). (See middle curve 451 inFIG. 33). Increasing the ceramic crystalline structure with longer heattreatment [30 min at 900°] produces even larger grain size (≅0.20micron, 200 nm), and higher values of relative permittivity, ∈_(R)≅1000.(See upper curve 453 in FIG. 33). However, larger grain size alsoproduces a thermal dependence in the relative permittivity that isincompatible with stable dielectric response versus temperature neededto make a high-Q antenna that is effective over the −40° C. to 85° C.temperature range. In a preferred embodiment, a high-κ ceramicinclusions that have a maximum nominal grain size ≦0.050 micron (50 nm)may be employed to obtain a composite dielectric body whose effectivepermittivity changes as a function of temperature ≦5×10⁻²° C.⁻¹. Inanother preferred embodiment, inclusions having a maximum nominal grainsize ≦0.035 micron (35 nm) can be incorporated into the compositemeta-material dielectric body, so as to obtain a composite dielectricbody whose effective permittivity changes as a function of temperature≦9×10⁻³° C.⁻¹. The smallest grain sizes used in powder-based aerosolsprays are on the order of 50 micron, therefore it is a preferred methodof the present invention to use liquid aerosol sprays to achieve greatercontrol over the high-κ ceramic microstructure, most importantly grainsize. The use of rapid thermal annealing techniques using laser,infrared light, or ultraviolet light sources provides greater controlover the energy delivered to the ceramic during the grain growth step.

The antenna modules and manufacturing methods described above have valuein any RF application that requires lightweight, compact size, and lowpower consumption, such as airborne radars, automotive radars, andmobile communications systems. The ceramic antenna modules describedherein have radiation patterns that protrude in the half-space thatcontains the antenna element. The precise radiation pattern is afunction of the shape of the antenna element, the number of antennaelements used to form the radiation pattern, as well as the preciselocation relative to the conducting antenna element and the relativedielectric properties of the ceramic inclusions embedded within a host.As shown in FIG. 34, these antenna modules can be used to developlow-profile customer premises equipment (CPE) radios useful in broadbandcommunications systems. Many of these systems will utilize signalingfrequencies that are greater than or equal to 2.4 GHz. These frequenciespropagate well through air and window glass, but can be sharplyattenuated by walls and other housing structures, which causes a problemfor last mile broadband wireless delivery systems. Many last milewireless systems require the use of large dish, or other antenna systemsthat are unattractive to the home consumer. The present inventionenables a CPE radio 455 solution incorporates a high gain ceramicantenna module 457 that has a narrowly focused radiation profile 459that is directed towards a network base station 461 transmission towerand is used to manage transceiver functions between the base station andthe customer premises. The low-profile CPE radio 455 is situated on awindow 463 with a view of the base station transmission tower 461 or astrong reflection point in non-line of sight (NLOS) locations. The CPEradio 455 also comprises a second antenna module 465 having a radiationpattern 467 that broadly disperses/detects signals within the customerpremises. The directional sensitivity of the ceramic antenna modulesenabled by the present invention also provides a means to improvewireless security as shown in FIGS. 35A&B. Wireless local area networksare often susceptible to security threats and unwanted networkintrusions because electromagnetic emissions 471 will extend beyond thecustomers' premises 473 exposing the network to security hackers locatednearby. A perimeter 475 of small form factor transceiver units 477comprising a receive antenna module 479 with one or more receive antennaelements is used to detect electromagnetic emissions 481 emanating fromwithin the premises 473. The electromagnetic emissions 481 are passedthrough an amplifier stage 483 that inverts the phase (shifts theirphase by 180 degrees) and rebroadcasts a phase-inverted electromagneticsignal 485 through a transmit antenna module 487 in the same directionof the electromagnetic emissions 481. A properly tuned amplifier stage483 will broadcast a phase-inverted signal with an amplitude such thatthe ratio between the primary signal and inverted signal is less than0.05 dB. Phase-cancellation between the network's electromagneticemission 487 and the phase-inverted signal 485 causes signals emittedfrom the customer premises 473 to be reduced to non-detectable limitsoutside the security perimeter 475.

The ability to reduce the physical size of an antenna element, or anarray of antenna elements, under the present invention enables efficientbroadband wireless communications in mobile platforms, such as cellularphones and laptop computers. This is particularly true for broadbandcommunications at the 400-800 MHz band, where resonant antenna lengthsfor the communications wavelengths (≈75 cm-37.5 cm) are larger than thedevices. FIGS. 36A&B depict the application of the antenna module in alaptop computer 489, where the antenna module 490A,B,C is situated inthe laptop body 491 adjacent to the key board 492, located either on atop 490A or side surface 490B. Antenna modules 493A,B, may alternativelybe situated on the back (top) side of the laptop body that contains thedisplay screen 494. Often it is desirable to have a second antennamodule 493B positioned nearby the first antenna module 493A to accountfor spatial diversity among the incoming signals. At sufficiently highfrequency the antenna pair may be engineered into the same module. Incell phone applications, it is more frequently desirable to have anantenna system that is omni-directional to account for movements on thepart of the user during operation. As shown in FIGS. 37A&B, it isdesirable to utilize two antenna modules 494A,B to provideomni-directional coverage as each will radiate/receive signals in thehalf-plane defined by the modules' ground plane. This is realized bylocating one antenna module 494A within the cellular phone device 495such that the module's transmit/receive radiation pattern is directed496 away from the surface 497 that contains the device's microphone 498and speaker 499. In most cellular phone devices this surface wouldcoincide with the surface that is placed in contact with the phoneusers' ear or face. The second antenna module is 494B positioned so itstransmit/receive radiation pattern is directed 500 away from theopposite surface 501. This antenna configuration further provides ameans to protect against the perception of human health hazards causedby microwave emissions by allowing the cell phone manufacturer to tunethe transmitted power from antenna module 494A to a lower level thanthat transmitted by antenna module 494B. Although the illustrationprovided by FIGS. 37A,B show the antenna modules 494A,B embedded in anexterior surface of the cell phone device, it should be understood thatthe modules 494A,B may also be situated below the surfaces 497,501.Reference is now made to FIG. 38 which depicts a satellite phone device502 where transmissions are directed skyward. In this embodiment, it ispreferred to locate the antenna modules 503 within a minor surface 504of the phone that is adjacent or opposite to the surface 505 thatcontains the phone's speaker 506 and microphone 507, and to project themodules' 503 transmit/receive radiation pattern in a skyward direction508 during normal operational use.

The small form factor ceramic antenna modules enabled by the presentinvention are thus useful in any appliance, a computer, printer, smartrefrigerator, etc., that will have a wireless interface. However, thesemodules may also be used in wireless appliance as shown in FIG. 39 byplacing the wireless antenna module 509A,B on or within a top (front)510 or side 511 surface of the wireless appliance 512.

Although the invention has been described with respect to variousembodiments, it should be realized this invention is also capable of awide variety of further and other embodiments within the spirit andscope of the appended claims.

What is claimed is:
 1. An antenna module, comprising: a compositedielectric body including at least one ceramic dielectric inclusioncomposed of at least one material selected from the group consisting ofmaterials having a relative dielectric permittivity ≧10 and materialshaving a relative dielectric permeability ≠1, the at least one ceramicdielectric inclusion embedded in a dielectric host material selectedfrom the group consisting of amorphous silica, titania, tantalates, purealumina, admixtures thereof, and an organic media such that thecomposite dielectric body has an effective permittivity ≧4; ametallization layer adjacent the composite dielectric body and includinga ground plane and at least one contact pad; and at least oneelectrically conductive element disposed parallel to the metallizationlayer and electrically connected with the at least one contact pad, eachconductive element disposed relative to the composite dielectric bodysuch that the conductive element is resonant over a band of frequenciesand has a length that is ≦50% of the length that would be required tomaintain the same resonance without the composite dielectric body,wherein the at least one electrically conductive element comprises aplurality of antenna elements.
 2. The module of claim 1, wherein each ofthe antenna elements is operable in a distinct frequency bandwidth. 3.An antenna module, comprising: a composite dielectric body including atleast one ceramic dielectric inclusion composed of at least one materialselected from the group consisting of materials having a relativedielectric permittivity ≧10 and materials having a relative dielectricpermeability ≠1, the at least one ceramic dielectric inclusion embeddedin a dielectric host material selected from the group consisting ofamorphous silica, titania, tantalates, pure alumina, admixtures thereof,and an organic media such that the composite dielectric body has aneffective permittivity ≧4; a metallization layer adjacent the compositedielectric body and including a ground plane and at least one contactpad; and at least one electrically conductive element disposed parallelto the metallization layer and electrically connected with the at leastone contact pad, each conductive element disposed relative to thecomposite dielectric body such that the conductive element is resonantover a band of frequencies and has a length that is ≦50% of the lengththat would be required to maintain the same resonance without thecomposite dielectric body, further comprising at least one capacitivepad upon which a corresponding one of the at least one electricallyconductive elements terminates, and wherein the at least one antennaelement is composed of discrete wire.
 4. The module of claim 3, whereinthe capacitive pad includes a landing portion having a width that isabout 2.5 um to 10 um wider than the cross-sectional diameter of thediscrete wire electrically conductive element that terminates upon it, alength sufficient to accommodate for variations in the discrete wirebonding process and a surface area ≦ about 5% of the surface area of theentire capacitive pad.